Systems and methods of RF power transmission, modulation, and amplification, including a Cartesian-Polar-Cartesian-Polar (CPCP) embodiment

ABSTRACT

Methods and systems for vector combining power amplification are disclosed herein. In one embodiment, a plurality of signals are individually amplified, then summed to form a desired time-varying complex envelope signal. Phase and/or frequency characteristics of one or more of the signals are controlled to provide the desired phase, frequency, and/or amplitude characteristics of the desired time-varying complex envelope signal. In another embodiment, a time-varying complex envelope signal is decomposed into a plurality of constant envelope constituent signals. The constituent signals are amplified equally or substantially equally, and then summed to construct an amplified version of the original time-varying envelope signal. Embodiments also perform frequency up-conversion.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. patent applicationSer. No. 11/256,172, filed Oct. 24, 2005, which claims the benefit ofU.S. Provisional Patent Application No. 60/620,972 filed on Oct. 22,2004, U.S. Provisional Patent Application No. 60/671,542 filed on Apr.15, 2005, U.S. Provisional Patent Application No. 60/671,536 filed onApr. 15, 2005, U.S. Provisional Patent Application No. 60/673,397 filedon Apr. 21, 2005, U.S. Provisional Patent Application No. 60/706,003filed on Aug. 8, 2005, U.S. Provisional Patent Application No.60/709,092 filed on Aug. 18, 2005, U.S. Provisional Patent ApplicationNo. 60/717,244 filed on Sep. 16, 2005, and U.S. Provisional PatentApplication No. 60/721,114 filed on Sep. 28, 2005, all of which areincorporated herein by reference in their entireties.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to modulation and on-frequencypower amplification. More particularly, the invention relates to methodsand systems for vector combining power amplification.

2. Background Art

In power amplifiers, a complex tradeoff typically exists betweenlinearity and power efficiency.

Linearity is determined by a power amplifier's operating range on acharacteristic curve that relates its input to output variables—the morelinear the operating range the more linear the power amplifier is saidto be. Linearity is a desired characteristic of a power amplifier. Inone aspect, for example, it is desired that a power amplifier uniformlyamplifies signals of varying amplitude, and/or phase and/or frequency.Accordingly, linearity is an important determiner of the output signalquality of a power amplifier.

Power efficiency can be calculated using the relationship of the totalpower delivered to a load divided by the total power supplied to theamplifier. For an ideal amplifier, power efficiency is 100%. Typically,power amplifiers are divided into classes which determine theamplifier's maximum theoretical power efficiency. Power efficiency isclearly a desired characteristic of a power amplifier—particularly, inwireless communication systems where power consumption is significantlydominated by the power amplifier.

Unfortunately, the traditional tradeoff between linearity and efficiencyin power amplifiers is such that the more linear a power amplifier isthe less power efficient it is. For example, the most linear amplifieris biased for class A operation, which is the least efficient class ofamplifiers. On the other hand, higher class amplifiers such as class B,C, D, E, etc, are more power efficient, but are considerably non-linearwhich can result in spectrally distorted output signals.

The tradeoff described above is further accentuated by typical wirelesscommunication signals. Wireless communication signals, such as OFDM,CDMA, and W-CDMA for example, are generally characterized by theirpeak-to-average power ratios. The larger the signal's peak to averageratio the more non-linear distortion will be produced when non-linearamplifiers are employed.

Outphasing amplification techniques have been proposed for RF amplifierdesigns. In several aspects, however, existing outphasing techniques aredeficient in satisfying complex signal amplification requirements,particularly as defined by wireless communication standards, forexample.

In one aspect, existing outphasing techniques employ an isolating and/ora combining element when combining constant envelope constituents of adesired output signal. For example, it is commonly the case that a powercombiner is used to combine the constituent signals. This combiningapproach, however, typically results in a degradation of output signalpower due to insertion loss and limited bandwidth, and, correspondingly,a decrease in power efficiency.

In another aspect, the typically large size of combining elementsprecludes having them in monolithic amplifier designs.

What is needed therefore are power amplification methods and systemsthat solve the deficiencies of existing power amplifying techniqueswhile maximizing power efficiency and minimizing non-linear distortion.Further, power amplification methods and systems that can be implementedwithout the limitations of traditional power combining circuitry andtechniques are needed.

BRIEF SUMMARY OF THE INVENTION

Embodiments for vector combining power amplification are disclosedherein.

In one embodiment, a plurality of substantially constant envelopesignals are individually amplified, then combined to form a desiredtime-varying complex envelope signal. Phase and/or frequencycharacteristics of one or more of the signals are controlled to providethe desired phase, frequency, and/or amplitude characteristics of thedesired time-varying complex envelope signal.

In another embodiment, a time-varying complex envelope signal isdecomposed into a plurality of substantially constant envelopeconstituent signals. The constituent signals are amplified, and thenre-combined to construct an amplified version of the originaltime-varying envelope signal.

Embodiments of the invention can be practiced with modulated carriersignals and with baseband information and clock signals. Embodiments ofthe invention also achieve frequency up-conversion. Accordingly,embodiments of the invention represent integrated solutions forfrequency up-conversion, amplification, and modulation.

Embodiments of the invention can be implemented with analog and/ordigital controls. The invention can be implemented with analogcomponents or with a combination of analog components and digitalcomponents. In the latter embodiment, digital signal processing can beimplemented in an existing baseband processor for added cost savings.

Additional features and advantages of the invention will be set forth inthe description that follows. Yet further features and advantages willbe apparent to a person skilled in the art based on the description setforth herein or may be learned by practice of the invention. Theadvantages of the invention will be realized and attained by thestructure and methods particularly pointed out in the writtendescription and claims hereof as well as the appended drawings.

It is to be understood that both the foregoing summary and the followingdetailed description are exemplary and explanatory and are intended toprovide further explanation of embodiments of the invention as claimed.

BRIEF DESCRIPTION OF THE FIGURES

Embodiments of the present invention will be described with reference tothe accompanying drawings, wherein generally like reference numbersindicate identical or functionally similar elements. Also, generally,the leftmost digit(s) of the reference numbers identify the drawings inwhich the associated elements are first introduced.

FIG. 1A is an example that illustrates the generation of an exemplarytime-varying complex envelope signal.

FIG. 1B is another example that illustrates the generation of anexemplary time-varying complex envelope signal.

FIG. 1C is an example that illustrates the generation of an exemplarytime-varying complex envelope signal from the sum of two or moreconstant envelope signals.

FIG. 1D illustrates the power amplification of an example time-varyingcomplex envelope signal according to an embodiment of the presentinvention.

FIG. 1E is a block diagram that illustrates a vector power amplificationembodiment of the present invention.

FIG. 1 illustrates a phasor representation of a signal.

FIG. 2 illustrates a phasor representation of a time-varying complexenvelope signal.

FIGS. 3A-3C illustrate an example modulation to generate a time-varyingcomplex envelope signal.

FIG. 3D is an example that illustrates constant envelope decompositionof a time-varying envelope signal.

FIG. 4 is a phasor diagram that illustrates a Cartesian 4-Branch VectorPower Amplification (VPA) method of an embodiment of the presentinvention.

FIG. 5 is a block diagram that illustrates an exemplary embodiment ofthe Cartesian 4-Branch VPA method.

FIG. 6 is a process flowchart embodiment for power amplificationaccording to the Cartesian 4-Branch VPA method.

FIG. 7A is a block diagram that illustrates an exemplary embodiment of avector power amplifier for implementing the Cartesian 4-Branch VPAmethod.

FIG. 7B is a block diagram that illustrates another exemplary embodimentof a vector power amplifier for implementing the Cartesian 4-Branch VPAmethod.

FIG. 8A is a block diagram that illustrates another exemplary embodimentof a vector power amplifier according to the Cartesian 4-Branch VPAmethod.

FIG. 8B is a block diagram that illustrates another exemplary embodimentof a vector power amplifier according to the Cartesian 4-Branch VPAmethod.

FIG. 8C is a block diagram that illustrates another exemplary embodimentof a vector power amplifier according to the Cartesian 4-Branch VPAmethod.

FIG. 8D is a block diagram that illustrates another exemplary embodimentof a vector power amplifier according to the Cartesian 4-Branch VPAmethod.

FIGS. 9A-9B are phasor diagrams that illustrate aCartesian-Polar-Cartesian-Polar (CPCP) 2-Branch Vector PowerAmplification (VPA) method of an embodiment of the present invention.

FIG. 10 is a block diagram that illustrates an exemplary embodiment ofthe CPCP 2-Branch VPA method.

FIG. 10A is a block diagram that illustrates another exemplaryembodiment of the CPCP 2-Branch VPA method.

FIG. 11 is a process flowchart embodiment for power amplificationaccording to the CPCP 2-Branch VPA method.

FIG. 12 is a block diagram that illustrates an exemplary embodiment of avector power amplifier for implementing the CPCP 2-Branch VPA method.

FIG. 12A is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the CPCP2-Branch VPA method.

FIG. 12B is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the CPCP2-Branch VPA method.

FIG. 13 is a block diagram that illustrates another exemplary embodimentof a vector power amplifier for implementing the CPCP 2-Branch VPAmethod.

FIG. 13A is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the CPCP2-Branch VPA method.

FIG. 14 is a phasor diagram that illustrates a Direct Cartesian 2-BranchVector Power Amplification (VPA) method of an embodiment of the presentinvention.

FIG. 15 is a block diagram that illustrates an exemplary embodiment ofthe Direct Cartesian 2-Branch VPA method.

FIG. 15A is a block diagram that illustrates another exemplaryembodiment of the Direct Cartesian 2-Branch VPA method.

FIG. 16 is a process flowchart embodiment for power amplificationaccording to the Direct Cartesian 2-Branch VPA method.

FIG. 17 is a block diagram that illustrates an exemplary embodiment of avector power amplifier for implementing the Direct Cartesian 2-BranchVPA method.

FIG. 17A is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the DirectCartesian 2-Branch VPA method.

FIG. 17B is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the DirectCartesian 2-Branch VPA method.

FIG. 18 is a block diagram that illustrates another exemplary embodimentof a vector power amplifier for implementing the Direct Cartesian2-Branch VPA method.

FIG. 18A is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier for implementing the DirectCartesian 2-Branch VPA method.

FIG. 19 is a process flowchart that illustrates an I and Q transferfunction embodiment according to the Cartesian 4-Branch VPA method.

FIG. 20 is a block diagram that illustrates an exemplary embodiment ofan I and Q transfer function according to the Cartesian 4-Branch VPAmethod.

FIG. 21 is a process flowchart that illustrates an I and Q transferfunction embodiment according to the CPCP 2-Branch VPA method.

FIG. 22 is a block diagram that illustrates an exemplary embodiment ofan I and Q transfer function according to the CPCP 2-Branch VPA method.

FIG. 23 is a process flowchart that illustrates an I and Q transferfunction embodiment according to the Direct Cartesian 2-Branch VPAmethod.

FIG. 24 is a block diagram that illustrates an exemplary embodiment ofan I and Q transfer function according to the Direct Cartesian 2-BranchVPA method.

FIG. 25 is a phasor diagram that illustrates the effect of waveformdistortion on a representation of a signal phasor.

FIG. 26 illustrates magnitude to phase transform functions according toan embodiment of the present invention.

FIG. 27 illustrates exemplary embodiments of biasing circuitry accordingto embodiments of the present invention.

FIG. 28 illustrates a method of combining constant envelope signalsaccording to an embodiment the present invention.

FIG. 29 illustrates a vector power amplifier output stage embodimentaccording to the present invention.

FIG. 30 is a block diagram of a power amplifier (PA) output stageembodiment.

FIG. 31 is a block diagram of another power amplifier (PA) output stageembodiment.

FIG. 32 is a block diagram of another power amplifier (PA) output stageembodiment.

FIG. 33 is a block diagram of another power amplifier (PA) output stageembodiment according to the present invention.

FIG. 34 is a block diagram of another power amplifier (PA) output stageembodiment according to the present invention.

FIG. 35 is a block diagram of another power amplifier (PA) output stageembodiment according to the present invention.

FIG. 36 is a block diagram of another power amplifier (PA) output stageembodiment according to the present invention.

FIG. 37 illustrates an example output signal according to an embodimentof the present invention.

FIG. 38 illustrates an exemplary PA embodiment.

FIG. 39 illustrates an example time-varying complex envelope PA outputsignal and a corresponding envelop signal.

FIG. 40 illustrates example timing diagrams of a PA output stagecurrent.

FIG. 41 illustrates exemplary output stage current control functions.

FIG. 42 is a block diagram of another power amplifier (PA) output stageembodiment.

FIG. 43 illustrates an exemplary PA stage embodiment.

FIG. 44 illustrates an exemplary waved-shaped PA output signal.

FIG. 45 illustrates a power control method.

FIG. 46 illustrates another power control method.

FIG. 47 illustrates an exemplary vector power amplifier embodiment.

FIG. 48 is a process flowchart for implementing output stage currentshaping according to an embodiment of the present invention.

FIG. 49 is a process flowchart for implementing harmonic controlaccording to an embodiment of the present invention.

FIG. 50 is a process flowchart for power amplification according to anembodiment of the present invention.

FIGS. 51A-I illustrate exemplary multiple-input single-output (MISO)output stage embodiments.

The present invention will be described with reference to theaccompanying drawings. The drawing in which an element first appears istypically indicated by the leftmost digit(s) in the correspondingreference number.

DETAILED DESCRIPTION OF THE INVENTION

Table of Contents

-   -   1. Introduction        -   1.1. Example Generation of Time-Varying Complex Envelope            Input Signals        -   1.2. Example Generation of Time-Varying Complex Envelope            Signals from Constant Envelope Signals        -   1.3. Vector Power Amplification Overview    -   2. General Mathematical Overview        -   2.1. Phasor Signal Representation        -   2.2. Time-Varying Complex Envelope Signals        -   2.3. Constant Envelope Decomposition of Time-Varying            Envelope Signals    -   3. Vector Power Amplification (VPA) Methods and Systems        -   3.1. Cartesian 4-Branch Vector Power Amplifier        -   3.2. Cartesian-Polar-Cartesian-Polar (CPCP) 2-Branch Vector            Power Amplifier        -   3.3. Direct Cartesian 2-Branch Vector Power Amplifier        -   3.4. I and Q Data to Vector Modulator Transfer Functions            -   3.4.1. Cartesian 4-Branch VPA Transfer Function            -   3.4.2. CPCP 2-Branch VPA Transfer Function            -   3.4.3. Direct Cartesian 2-Branch VPA Transfer Function            -   3.4.4. Magnitude to Phase Shift Transform                -   3.4.4.1. Magnitude to Phase Shift Transform for                    Sinusoidal Signals                -   3.4.4.2. Magnitude to Phase Shift Transform for                    Square Wave Signals            -   3.4.5. Waveform Distortion Compensation        -   3.5. Output Stage            -   3.5.1. Output Stage Embodiments            -   3.5.2. Output Stage Current Shaping            -   3.5.3. Output Stage Protection        -   3.6. Harmonic Control        -   3.7. Power Control        -   3.8. Exemplary Vector Power Amplifier Embodiment    -   4. Summary    -   5. Conclusions

1. Introduction

Methods, apparatuses and systems for vector combining poweramplification are disclosed herein.

Vector combining power amplification is an approach for optimizinglinearity and power efficiency simultaneously. Generally speaking, andreferring to flowchart 502 in FIG. 50, in step 504 a time-varyingcomplex envelope input signal, with varying amplitude and phase, isdecomposed into constant envelope constituent signals. In step 506, theconstant envelope constituent signals are amplified, and then in step508 summed to generate an amplified version of the input complexenvelope signal. Since substantially constant envelope signals may beamplified with minimal concern for non-linear distortion, the result ofsumming the constant envelope signals suffers minimal non-lineardistortion while providing optimum efficiency.

Accordingly, vector combining power amplification allows for non-linearpower amplifiers to be used to efficiently amplify complex signalswhilst maintaining minimal non-linear distortion levels.

For purposes of convenience, and not limitation, methods and systems ofthe present invention are sometimes referred to herein as vector poweramplification (VPA) methods and systems.

A high-level description of VPA methods and systems according toembodiments of the present invention is now provided. For the purpose ofclarity, certain terms are first defined below. The definitionsdescribed in this section are provided for convenience purposes only,and are not limiting. The meaning of these terms will be apparent topersons skilled in the art(s) based on the entirety of the teachingsprovided herein. These terms may be discussed throughout thespecification with additional detail.

The term signal envelope, when used herein, refers to an amplitudeboundary within which a signal is contained as it fluctuates in the timedomain. Quadrature-modulated signals can be described by

r(t) = i(t) ⋅ cos (ω c ⋅ t) + q(t) ⋅ sin (ω c ⋅ t)where i(t) and q(t) represent in-phase and quadrature signals with thesignal envelope e(t), being equal to e(t)=√{square root over(i(t)²+q(t)²)}{square root over (i(t)²+q(t)²)} and the phase angleassociated with r(t) is related to arctan (q(t)/i(t).

The term constant envelope signal, when used herein, refers to in-phaseand quadrature signals where e(t)=√{square root over(i(t)²+q(t)²)}{square root over (i(t)²+q(t)²)}, with e(t) having arelatively or substantially constant value.

The term time-varying envelope signal, when used herein, refers to asignal having a time-varying signal envelope. A time-varying envelopesignal can be described in terms of in-phase and quadrature signals ase(t)=√{square root over (i(t)²+q(t)²)}{square root over (i(t)²+q(t)²)},with e(t) having a time-varying value.

The term phase shifting, when used herein, refers to delaying oradvancing the phase component of a time-varying or constant envelopesignal relative to a reference phase.

1.1) Example Generation of Complex Envelope Time-Varying Input Signals

FIGS. 1A and 1B are examples that illustrate the generation oftime-varying envelope and phase complex input signals. In FIG. 1A,time-varying envelope carrier signals 104 and 106 are input into phasecontroller 110. Phase controller 110 manipulates the phase components ofsignals 104 and 106. In other words, phase controller 110 may phaseshift signals 104 and 106. Resulting signals 108 and 112, accordingly,may be phased shifted relative to signals 104 and 106. In the example ofFIG. 1A, phase controller 110 causes a phase reversal (180 degree phaseshift) in signals 104 and 106 at time instant t_(o), as can be seen fromsignals 108 and 112. Signals 108 and 112 represent time-varying complexcarrier signals. Signals 108 and 112 have both time-varying envelopesand phase components. When summed, signals 108 and 112 result in signal114. Signal 114 also represents a time-varying complex signal. Signal114 may be an example input signal into VPA embodiments of the presentinvention (for example, an example input into step 504 of FIG. 50).

Time-varying complex signals may also be generated as illustrated inFIG. 1B. In FIG. 1B, signals 116 and 118 represent baseband signals. Forexample, signals 116 and 118 may be in-phase (I) and quadrature (Q)baseband components of a signal. In the example of FIG. 1B, signals 116and 118 undergo a zero crossing as they transition from +1 to −1.Signals 116 and 118 are multiplied by signal 120 or signal 120 phaseshifted by 90 degrees. Signal 116 is multiplied by a 0 degree shiftedversion of signal 120. Signal 118 is multiplied by a 90 degree shiftedversion of signal 120. Resulting signals 122 and 124 representtime-varying complex carrier signals. Note that signals 122 and 124 haveenvelopes that vary according to the time-varying amplitudes of signals116 and 118. Further, signals 122 and 124 both undergo phase reversalsat the zero crossings of signals 116 and 118. Signals 122 and 124 aresummed to result in signal 126. Signal 126 represents a time-varyingcomplex signal. Signal 126 may represent an example input signal intoVPA embodiments of the present invention. Additionally, signals 116 and118 may represent example input signals into VPA embodiments of thepresent invention.

1.2) Example Generation of Time-Varying Complex Envelope Signals fromConstant Envelope Signals

The description in this section generally relates to the operation ofstep 508 in FIG. 50. FIG. 1C illustrates three examples for thegeneration of time-varying complex signals from the sum of two or moresubstantially constant envelope signals. A person skilled in the artwill appreciate, however, based on the teachings provided herein thatthe concepts illustrated in the examples of FIG. 1C can be similarlyextended to the case of more than two constant envelope signals.

In example 1 of FIG. 1C, constant envelope signals 132 and 134 are inputinto phase controller 130. Phase controller 130 manipulates phasecomponents of signals 132 and 134 to generate signals 136 and 138,respectively. Signals 136 and 138 represent substantially constantenvelope signals, and are summed to generate signal 140. The phasorrepresentation in FIG. 1C, associated with example 1 illustrates signals136 and 138 as phasors P₁₃₆ and P₁₃₈, respectively. Signal 140 isillustrated as phasor P₁₄₀. In example 1, P₁₃₆ and P₁₃₈ aresymmetrically phase shifted by an angle φ₁ relative to a referencesignal assumed to be aligned with the real axis of the phasorrepresentation. Correspondingly, time domain signals 136 and 138 arephase shifted in equal amounts but opposite directions relative to thereference signal. Accordingly, P₁₄₀, which is the sum of P₁₃₆ and P₁₃₈,is in-phase with the reference signal.

In example 2 of FIG. 1C, substantially constant envelope signals 132 and134 are input into phase controller 130. Phase controller 130manipulates phase components of signals 132 and 134 to generate signals142 and 144, respectively. Signals 142 and 144 are substantiallyconstant envelope signals, and are summed to generate signal 150. Thephasor representation associated with example 2 illustrates signals 142and 144 as phasors P₁₄₂ and P₁₄₄, respectively. Signal 150 isillustrated as phasor P₁₅₀. In example 2, P₁₄₂ and P₁₄₄ aresymmetrically phase shifted relative to a reference signal. Accordingly,similar to P₁₄₀, P₁₅₀ is also in-phase with the reference signal. P₁₄₂and P₁₄₄, however, are phase shifted by an angle whereby φ₂≠φ₁ relativeto the reference signal. P₁₅₀, as a result, has a different magnitudethan P₁₄₀ of example 1. In the time domain representation, it is notedthat signals 140 and 150 are in-phase but have different amplitudesrelative to each other.

In example 3 of FIG. 1C, substantially constant envelope signals 132 and134 are input into phase controller 130. Phase controller 130manipulates phase components of signals 132 and 134 to generate signals146 and 148, respectively. Signals 146 and 148 are substantiallyconstant envelope signals, and are summed to generate signal 160. Thephasor representation associated with example 3 illustrates signals 146and 148 as phasors P₁₄₆ and P₁₄₈, respectively. Signal 160 isillustrated as phasor P₁₆₀. In example 3, P₁₄₆ is phased shifted by anangle φ₃ relative to the reference signal. P₁₄₈ is phase shifted by anangle φ₄ relative to the reference signal. φ₃ and φ₄ may or may not beequal. Accordingly, P₁₆₀, which is the sum of P₁₄₆ and P₁₄₈, is nolonger in-phase with the reference signal. P₁₆₀ is phased shifted by anangle Θ relative to the reference signal. Similarly, P₁₆₀ is phaseshifted by Θ relative to P₁₄₀ and P₁₅₀ of examples 1 and 2. P₁₆₀ mayalso vary in amplitude relative to P₁₄₀ as illustrated in example 3.

In summary, the examples of FIG. 1C demonstrate that a time-varyingamplitude signal can be obtained by the sum of two or more substantiallyconstant envelope signals (Example 1). Further, the time-varying signalcan have amplitude changes but no phase changes imparted thereon byequally shifting in opposite directions the two or more substantiallyconstant envelope signals (Example 2). Equally shifting in the samedirection the two or more constant envelope constituents of the signal,phase changes but no amplitude changes can be imparted on thetime-varying signal. Any time-varying amplitude and phase signal can begenerated using two or more substantially constant envelope signals(Example 3).

It is noted that signals in the examples of FIG. 1C are shown assinusoidal waveforms for purpose of illustration only. A person skilledin the art will appreciate based on the teachings herein that othertypes of waveforms may also have been used. It should also be noted thatthe examples of FIG. 1C are provided herein for the purpose ofillustration only, and may or may not correspond to a particularembodiment of the present invention.

1.3) Vector Power Amplification Overview

A high-level overview of vector power amplification is now provided.FIG. 1D illustrates the power amplification of an exemplary time-varyingcomplex input signal 172. Signals 114 and 126 as illustrated in FIGS. 1Aand 1B may be examples of signal 172. Further, signal 172 may begenerated by or comprised of two or more constituent signals such as 104and 106 (FIG. 1A), 108 and 112 (FIG. 1A), 116 and 118 (FIG. 1B), and 122and 124 (FIG. 1B).

In the example of FIG. 1D, VPA 170 represents a VPA system embodimentaccording to the present invention. VPA 170 amplifies signal 172 togenerate amplified output signal 178. Output signal 178 is amplifiedefficiently with minimal distortion.

In the example of FIG. 1D, signals 172 and 178 represent voltage signalsV_(in)(t) and V_(olt)(t), respectively. At any time instant, in theexample of FIG. 1D, V_(in)(t) and V_(olt)(t) are related such thatV_(olt)(t)=Kev_(in)(tat′), where K is a scale factor and t′ represents atime delay that may be present in the VPA system. For power implication,

${\frac{V_{out}^{2}(t)}{Z_{out}} > \frac{V_{in}^{2}(t)}{Z_{in}}},$where output signal 178 is a power amplified version of input signal172.

Linear (or substantially linear) power amplification of time-varyingcomplex signals, as illustrated in FIG. 1D, is achieved according toembodiments of the present as shown in FIG. 1E.

FIG. 1E is an example block diagram that conceptually illustrates avector power amplification embodiment according to embodiments of thepresent invention. In FIG. 1E, input signal 172 represents atime-varying complex signal. For example, input signal 172 may begenerated as illustrated in FIGS. 1A and 1B. In embodiments, signal 172may be a digital or an analog signal. Further, signal 172 may be abaseband or a carrier-based signal.

Referring to FIG. 1E, according to embodiments of the present invention,input signal 172 or equivalents thereof are input into VPA 182. In theembodiment of FIG. 1E, VPA 182 includes a state machine 184 and analogcircuitry 186. State machine 184 may include digital and/or analogcomponents. Analog circuitry 186 includes analog components. VPA 182processes input signal 172 to generate two or more signals 188-{1, . . ., n}, as illustrated in FIG. 1E. As described with respect to signals136, 138, 142, 144, and 146, 148, in FIG. 1C, signals 188-{1, . . . , n}may or may not be phase shifted relative to each other over differentperiods of time. Further, VPA 182 generates signals 188-{1, . . . , n}such that a sum of signals 188-{1, . . . , n} results in signal 194which, in certain embodiments, can be an amplified version of signal172.

Still referring to FIG. 1E, signals 188-{1, . . . , n} are substantiallyconstant envelope signals. Accordingly, the description in the priorparagraph corresponds to step 504 in FIG. 50.

In the example of FIG. 1E, generally corresponding to step 506 in FIG.50, constant envelope signals 188-{1, . . . , n} are each independentlyamplified by a corresponding power amplifier (PA) 190-{1, . . . , f} togenerate amplified signals 192-{1, . . . , n}. In embodiments, PAs190-{1, . . . , n} amplify substantially equally respective constantenvelope signals 188-{1, . . . , n}. Amplified signals 192-{1, . . . ,n} are substantially constant envelope signals, and in step 508 aresummed to generate output signal 194. Note that output signal 194 can bea linearly (or substantially linearly) amplified version of input signal172. Output signal 194 may also be a frequency-upconverted version ofinput signal 172, as described herein.

2. General Mathematical Overview

2.1) Phasor Signal Representation

FIG. 1 illustrates a phasor representation {right arrow over (R)} 102 ofa signal r(t). A phasor representation of a signal is explicitlyrepresentative of the magnitude of the signal's envelope and of thesignal's phase shift relative to a reference signal. In this document,for purposes of convenience, and not limitation, the reference signal isdefined as being aligned with the real (Re) axis of the orthogonal spaceof the phasor representation. The invention is not, however, limited tothis embodiment. The frequency information of the signal is implicit inthe representation, and is given by the frequency of the referencesignal. For example, referring to FIG. 1, and assuming that the realaxis corresponds to a cos(ωt) reference signal, phasor {right arrow over(R)} would translate to the function r(t)=dR(t)cos(ωt+φ(t)), where R isthe magnitude of {right arrow over (R)}.

Still referring to FIG. 1, it is noted that phasor {right arrow over(R)} can be decomposed into a real part phasor {right arrow over (I)}and an imaginary part phasor {right arrow over (Q)}. {right arrow over(I)} and {right arrow over (Q)} are said to be the in-phase andquadrature phasor components of {right arrow over (R)} with respect tothe reference signal. It is further noted that the signals thatcorrespond to {right arrow over (I)} and {right arrow over (Q)} arerelated to r(t) as I(t)=R(t)·cos(φ(t)) and Q(t)=R(t)·sin(φ(t)),respectively. In the time domain, signal r(t) can also be written interms of its in-phase and quadrature components as follows:r(t)=I(t)·cos(ωt)+Q(t)·sin(ωt)=R(t)·cos(φ(t))·cos(ωt)+R(t)·sin(φ(t))·sin(ωt)  (1)

Note that, in the example of FIG. 1, R(t) is illustrated at a particularinstant of time.

2.2) Time-Varying Complex Envelope Signals

FIG. 2 illustrates a phasor representation of a signal r(t) at twodifferent instants of time t1 and t2. It is noted that the magnitude ofthe phasor, which represents the magnitude of the signal's envelope, aswell as its relative phase shift both vary from time t1 to time t2. InFIG. 2, this is illustrated by the varying magnitude of phasors {rightarrow over (R₁)} and {right arrow over (R₂)} and their correspondingphase shift angles φ₁ and φ₂. Signal r(t), accordingly, is atime-varying complex envelope signal.

It is further noted, from FIG. 2, that the real and imaginary phasorcomponents of signal r(t) are also time-varying in amplitude.Accordingly, their corresponding time domain signals also havetime-varying envelopes.

FIGS. 3A-3C illustrate an example modulation to generate a time-varyingcomplex envelope signal. FIG. 3A illustrates a view of a signal m(t).FIG. 3B illustrates a view of a portion of a carrier signal c(t). FIG.3C illustrates a signal r(t) that results from the multiplication ofsignals m(t) and c(t).

In the example of FIG. 3A, signal m(t) is a time-varying magnitudesignal. m(t) further undergoes a zero crossing. Carrier signal c(t), inthe example of FIG. 3B, oscillates at some carrier frequency, typicallyhigher than that of signal m(t).

From FIG. 3C, it can be noted that the resulting signal r(t) has atime-varying envelope. Further, it is noted, from FIG. 3C, that r(t)undergoes a reversal in phase at the moment when the modulating signalm(t) crosses zero. Having both non-constant envelope and phase, r(t) issaid to be a time-varying complex envelope signal.

2.3) Constant Envelope Decomposition of Time-Varying Envelope Signals

Any phasor of time-varying magnitude and phase can be obtained by thesum of two or more constant magnitude phasors having appropriatelyspecified phase shifts relative to a reference phasor.

FIG. 3D illustrates a view of an example time-varying envelope and phasesignal S(t). For ease of illustration, signal S(t) is assumed to be asinusoidal signal having a maximum envelope magnitude A. FIG. 3D furthershows an example of how signal S(t) can be obtained, at any instant oftime, by the sum of two constant envelope signals S₁(t) and S₂(t).Generally, S₁(t)=A₁ sin(ωt+φ₁(t)) and S₁(t)=A₂ sin(ωt+φ₂(t)).

For the purpose of illustration, three views are provided in FIG. 3Dthat illustrate how by appropriately phasing signals S₁(t) and S₂(t)relative to S(t), signals S₁(t) and S₂(t) can be summed so thatS(t)=K(S₁(t)+S₂(t)) where K is a constant. In other words, signal S(t)can be decomposed, at any time instant, into two or more signals. FromFIG. 3D, over period T₁, S₁(t) and S₂(t) are both in-phase relative tosignal S(t), and thus sum to the maximum envelope magnitude A of signalS(t). Over period T₃, however, signals S₁(t) and S₂(t) are 180 degreeout-of-phase relative to each other, and thus sum to a minimum envelopemagnitude of signal S(t).

The example of FIG. 3D illustrates the case of sinusoidal signals. Aperson skilled in the art, however, will understand that anytime-varying envelope, which modulates a carrier signal that can berepresented by a Fourier series or Fourier transform, can be similarlydecomposed into two or more substantially constant envelope signals.Thus, by controlling the phase of a plurality of substantially constantenvelope signals, any time-varying complex envelope signal can begenerated.

3. Vector Power Amplification Methods and Systems

Vector power amplification methods and systems according to embodimentsof the present invention rely on the ability to decompose anytime-varying envelope signal into two or more substantially constantenvelope constituent signals or to receive or generate such constituentsignals, amplify the constituent signals, and then sum the amplifiedsignals to generate an amplified version of the time-varying complexenvelope signal.

In sections 3.1-3.3, vector power amplification (VPA) embodiments of thepresent invention are provided, including 4-branch and 2-branchembodiments. In the description, each VPA embodiment is first presentedconceptually using a mathematical derivation of underlying concepts ofthe embodiment. An embodiment of a method of operation of the VPAembodiment is then presented, followed by various system levelembodiments of the VPA embodiment.

Section 3.4 presents various embodiments of control modules according toembodiments of the present invention. Control modules according toembodiments of the present invention may be used to enable certain VPAembodiments of the present invention. In some embodiments, the controlmodules are intermediary between an input stage of the VPA embodimentand a subsequent vector modulation stage of the VPA embodiment.

Section 3.5 describes VPA output stage embodiments according toembodiments of the present invention. Output stage embodiments aredirected to generating the output signal of a VPA embodiment.

Section 3.6 is directed to harmonic control according to embodiments ofthe present invention. Harmonic control may be implemented in certainembodiments of the present invention to manipulate the real andimaginary power in the harmonics of the VPA embodiment, thus increasingthe power present in the fundamental frequency at the output.

Section 3.7 is directed to power control according to embodiments of thepresent invention. Power control may be implemented in certainembodiments of the present invention in order to satisfy power levelrequirements of applications where VPA embodiments of the presentinvention may be employed.

3.1) Cartesian 4-Branch Vector Power Amplifier

According to one embodiment of the invention, herein called theCartesian 4-Branch VPA embodiment for ease of illustration and notlimitation, a time-varying complex envelope signal is decomposed into 4substantially constant envelope constituent signals. The constituentsignals are equally or substantially equally amplified individually, andthen summed to construct an amplified version of the originaltime-varying complex envelope signal.

It is noted that 4 branches are employed in this embodiment for purposesof illustration, and not limitation. The scope of the invention coversuse of other numbers of branches, and implementation of such variationswill be apparent to persons skilled in the art based on the teachingscontained herein.

In one embodiment, a time-varying complex envelope signal is firstdecomposed into its in-phase and quadrature vector components. In phasorrepresentation, the in-phase and quadrature vector components correspondto the signal's real part and imaginary part phasors, respectively.

As described above, magnitudes of the in-phase and quadrature vectorcomponents of a signal vary proportionally to the signal's magnitude,and are thus not constant envelope when the signal is a time-varyingenvelope signal. Accordingly, the 4-Branch VPA embodiment furtherdecomposes each of the in-phase and quadrature vector components of thesignal into four substantially constant envelope components, two for thein-phase and two for the quadrature signal components. This concept isillustrated in FIG. 4 using a phasor signal representation.

In the example of FIG. 4, phasors {right arrow over (I₁)} and {rightarrow over (I₂)} correspond to the real part phasors of an exemplarytime-varying complex envelope signal at two instants of time t1 and t2,respectively. It is noted that phasors {right arrow over (I₁)} and{right arrow over (I₂)} have different magnitudes.

Still referring to FIG. 4, at instant t1, phasor {right arrow over (I₁)}can be obtained by the sum of upper and lower phasors {right arrow over(I_(U) ₁ )} and {right arrow over (I_(L) ₁ )}. Similarly, at instant t2,phasor {right arrow over (I₂)} can be obtained by the sum of upper andlower phasors {right arrow over (I_(U) ₂ )} and {right arrow over (I_(L)₂ )}. Note that phasors {right arrow over (I_(U) ₁ )} and {right arrowover (I_(U) ₂ )} have equal or substantially equal magnitude. Similarly,phasors {right arrow over (I_(L) ₁ )} and {right arrow over (I_(L) ₂ )}have substantially equal magnitude. Accordingly, the real part phasor ofthe time-varying envelope signal can be obtained at any time instant bythe sum of at least two substantially constant envelope components.

The phase shifts of phasors {right arrow over (I_(U) ₁ )} and {rightarrow over (I_(L) ₁ )} relative to {right arrow over (I₁)}, as well asthe phase shifts of phasors {right arrow over (I_(U) ₂ )} and {rightarrow over (I_(L) ₂ )} relative to {right arrow over (I₂)} are setaccording to the desired magnitude of phasors {right arrow over (I₁)}and {right arrow over (I₂)}, respectively. In one case, when the upperand lower phasors are selected to have equal magnitude, the upper andlower phasors are symmetrically shifted in phase relative to the phasor.This is illustrated in the example of FIG. 4, and corresponds to {rightarrow over (I_(U) ₁ )}, {right arrow over (I_(L) ₁ )}, {right arrow over(I_(U) ₂ )}, and {right arrow over (I_(L) ₂ )} all having equalmagnitude. In a second case, the phase shift of the upper and lowerphasors are substantially symmetrically shifted in phase relative to thephasor. Based on the description herein, anyone skilled in the art willunderstand that the magnitude and phase shift of the upper and lowerphasors do not have to be exactly equal in value

As an example, it can be further verified that, for the case illustratedin FIG. 4, the relative phase shifts, illustrated as

$\frac{\phi_{1}}{2}$ and $\frac{\phi_{2}}{2}$in FIG. 4, are related to the magnitudes of normalized phasors {rightarrow over (I₁)} and {right arrow over (I₂)} as follows:

$\begin{matrix}{{{\frac{\phi_{1}}{2}\; = \;{\cot^{- 1}\left( \frac{I_{1}}{2\;\sqrt{\;{1\; - \;\frac{I_{1}^{2}}{4}}}} \right)}};}{and}} & (2) \\{{\frac{\phi_{2}}{2} = {\cot^{- 1}\left( \frac{I_{2}}{2\sqrt{1 - \frac{I_{2}^{2}}{4}}} \right)}},} & (3)\end{matrix}$

wherein I₁ and I₂ represent the normalized magnitudes of phasors {rightarrow over (I₁)} and {right arrow over (I₂)}, respectively, and whereinthe domains of I₁ and I₂ are restricted appropriately according to thedomain over which equation (2) and (3) are valid. It is noted thatequations (2) and (3) are one representation for relating the relativephase shifts to the normalized magnitudes. Other, solutions, equivalentrepresentations, and/or simplified representations of equations (2) and(3) may also be employed. Look up tables relating relative phase shiftsto normalized magnitudes may also be used.

The concept describe above can be similarly applied to the imaginaryphasor or the quadrature component part of a signal r(t) as illustratedin FIG. 4. Accordingly, at any time instant t, imaginary phasor part{right arrow over (Q)} of signal r(t) can be obtained by summing upperand lower phasor components {right arrow over (Q_(U))} and {right arrowover (Q_(L))} of substantially equal and constant magnitude. In thisexample, {right arrow over (Q_(U))} and {right arrow over (Q_(L))} aresymmetrically shifted in phase relative to {right arrow over (Q)} by anangle set according to the magnitude of {right arrow over (Q)} at timet. The relationship of {right arrow over (Q_(U))} and {right arrow over(Q_(L))} to the desired phasor {right arrow over (Q)} are related asdefined in equations 2 and 3 by substituting Q₁ and Q₂ for I₁ and I₂respectively.

It follows from the above discussion that, in phasor representation, anyphasor {right arrow over (R)} of variable magnitude and phase can beconstructed by the sum of four substantially constant magnitude phasorcomponents:{right arrow over (R)}={right arrow over (I _(U))}+{right arrow over (I_(L))}+{right arrow over (Q _(U))}+{right arrow over (Q _(L))};{right arrow over (I _(U))}+{right arrow over (I _(L))}={right arrowover (I)};{right arrow over (Q _(U))}+{right arrow over (Q _(L))}={right arrowover (Q)};I _(U) =I _(L)=constant;Q _(U) =Q _(L) =constant;  (4)where I_(U), I_(L), Q_(U), and Q_(L) represent the magnitudes of phasors{right arrow over (I_(U))}, {right arrow over (I_(L))}, {right arrowover (Q_(U))}, and {right arrow over (Q_(L))}, respectively.

Correspondingly, in the time domain, a time-varying complex envelopesinusoidal signal r(t)=R(t)cos(ωt+φ) is constructed by the sum of fourconstant envelope signals as follows:

$\begin{matrix}{{{{r(t)} = {{I_{U}(t)} + {I_{L}(t)} + {Q_{U}(t)} + {Q_{L}(t)}}};}{{{I_{U}(t)} = {{{{sgn}\left( \overset{->}{I} \right)} \times I_{U} \times {\cos\left( \frac{\phi_{I}}{2} \right)} \times {\cos\left( {\omega\; t} \right)}} + {I_{U} \times {\sin\left( \frac{\phi_{I}}{2} \right)} \times {\sin\left( {\omega\; t} \right)}}}};}{{{I_{L}(t)} = {{{{sgn}\left( \overset{->}{I} \right)} \times I_{L} \times {\cos\left( \frac{\phi_{I}}{2} \right)} \times {\cos\left( {\omega\; t} \right)}} - {I_{L} \times {\sin\left( \frac{\phi_{I}}{2} \right)} \times {\sin\left( {\omega\; t} \right)}}}};}{{{Q_{U}(t)} = {{{- {{sgn}\left( \overset{->}{Q} \right)}} \times Q_{U} \times {\cos\left( \frac{\phi_{Q}}{2} \right)} \times {\sin\left( {\omega\; t} \right)}} + {Q_{U} \times {\sin\left( \frac{\phi_{Q}}{2} \right)} \times {\cos\left( {\omega\; t} \right)}}}};}{{Q_{L}(t)} = {{{- {{sgn}\left( \overset{->}{Q} \right)}} \times Q_{L} \times {\cos\left( \frac{\phi_{Q}}{2} \right)} \times {\sin\left( {\omega\; t} \right)}} - {Q_{L} \times {\sin\left( \frac{\phi_{Q}}{2} \right)} \times {{\cos\left( {\omega\; t} \right)}.}}}}} & (5)\end{matrix}$where sgn({right arrow over (I)})=±1 depending on whether {right arrowover (I)} is in-phase or 180° degrees out-of-phase with the positivereal axis. Similarly, sgn({right arrow over (Q)})=±1 depending onwhether {right arrow over (Q)} is in-phase or 180° degrees out-of-phasewith the imaginary axis.

$\frac{\phi_{I}}{2}$corresponds to the phase shift of {right arrow over (I_(U))} and {rightarrow over (I_(L))} relative to the real axis. Similarly,

$\frac{\phi_{Q}}{2}$corresponds to the phase shift of {right arrow over (Q_(U))} and {rightarrow over (Q_(L))} relative to the imaginary axis.

$\frac{\phi_{I}}{2}$ and $\frac{\phi_{Q}}{2}$can be calculated using the equations given in (2) and (3).

Equations (5) can be further simplified as:

$\begin{matrix}{{{{r(t)} = {{I_{U}(t)} + {I_{L}(t)} + {Q_{U}(t)} + {Q_{L}(t)}}};}{{{I_{U}(t)} = {{{{sgn}\left( \overset{->}{I} \right)} \times I_{UX} \times {\cos\left( {\omega\; t} \right)}} + {I_{UY} \times {\sin\left( {\omega\; t} \right)}}}};}{{{I_{L}(t)} = {{{{sgn}\left( \overset{->}{I} \right)} \times I_{UX} \times {\cos\left( {\omega\; t} \right)}} - {I_{UY} \times {\sin\left( {\omega\; t} \right)}}}};}{{{Q_{U}(t)} = {{{- Q_{UY}} \times {\cos\left( {\omega\; t} \right)}} + {{{sgn}\left( \overset{->}{Q} \right)} \times Q_{UY} \times {\sin\left( {\omega\; t} \right)}}}};}{{Q_{L}(t)} = {{Q_{UY} \times {\cos\left( {\omega\; t} \right)}} - {{{sgn}\left( \overset{->}{Q} \right)} \times Q_{UY} \times {{\sin\left( {\omega\; t} \right)}.{where}}}}}{{I_{UX} = {{I_{U} \times {\cos\left( \frac{\phi_{I}}{2} \right)}} = {I_{L} \times {\cos\left( \frac{\phi_{Q}}{2} \right)}}}},{I_{UY} = {{I_{U} \times {\sin\left( \frac{\phi_{I}}{2} \right)}} = {I_{L} \times {\sin\left( \frac{\phi_{I}}{2} \right)}}}},{Q_{UX} = {{Q_{U} \times {\sin\left( \frac{\phi_{Q}}{2} \right)}} = {Q_{L} \times {\sin\left( \frac{\phi_{Q}}{2} \right)}}}},{and}}\text{}{Q_{UY} = {{Q_{U} \times {\cos\left( \frac{\phi_{Q}}{2} \right)}} = {Q_{L} \times {{\cos\left( \frac{\phi_{Q}}{2} \right)}.}}}}} & (6)\end{matrix}$

It can be understood by a person skilled in the art that, whereas thetime domain representations in equations (5) and (6) have been providedfor the case of a sinusoidal waveform, equivalent representations can bedeveloped for non-sinusoidal waveforms using appropriate basisfunctions. Further, as understood by a person skilled in the art basedon the teachings herein, the above-describe two-dimensionaldecomposition into substantially constant envelope signals can beextended appropriately into a multi-dimensional decomposition.

FIG. 5 is an example block diagram of the Cartesian 4-Branch VPAembodiment. An output signal r(t) 578 of desired power level andfrequency characteristics is generated from baseband in-phase andquadrature components according to the Cartesian 4-Branch VPAembodiment.

In the example of FIG. 5, a frequency generator such as a synthesizer510 generates a reference signal A*cos(ωt) 511 having the same frequencyas that of output signal r(t) 578. It can be understood by a personskilled in the art that the choice of the reference signal is madeaccording to the desired output signal. For example, if the desiredfrequency of the desired output signal is 2.4 GHz, then the frequency ofthe reference signal is set to be 2.4 GHz. In this manner, embodimentsof the invention achieve frequency up-conversion.

Referring to FIG. 5, one or more phase splitters are used to generatesignals 521, 531, 541, and 551 based on the reference signal 511. In theexample of FIG. 5, this is done using phase splitters 512, 514, and 516and by applying 0° phase shifts at each of the phase splitters. A personskilled in the art will appreciate, however, that various techniques maybe used for generating signals 521, 531, 541, and 551 of the referencesignal 511. For example, a 1:4 phase splitter may be used to generatethe four replicas 521, 531, 541, and 551 in a single step or in theexample embodiment of FIG. 5, signal 511 can be directly coupled tosignals 521, 531, 541, 551 Depending on the embodiment, a variety ofphase shifts may also be applied to result in the desired signals 521,531, 541, and 551.

Still referring to FIG. 5, the signals 521, 531, 541, and 551 are eachprovided to a corresponding vector modulator 520, 530, 540, and 550,respectively. Vector modulators 520, 530, 540, and 550, in conjunctionwith their appropriate input signals, generate four constant envelopeconstituents of signal r(t) according to the equations provided in (6).In the example embodiment of FIG. 5, vector modulators 520 and 530generate the I_(U)(t) and I_(L)(t) components, respectively, of signalr(t). Similarly, vector modulators 540 and 550 generate the Q_(U)(t) andQ_(L)(t) components, respectively, of signal r(t).

The actual implementation of each of vector modulators 520, 530, 540,and 550 may vary. It will be understood by a person skilled in the art,for example, that various techniques exist for generating the constantenvelope constituents according to the equations in (6).

In the example embodiment of FIG. 5, each of vector modulators 520, 530,540, 550 includes an input phase splitter 522, 532, 542, 552 for phasingthe signals 522, 531, 541, 551. Accordingly, input phase splitters 522,532, 542, 552 are used to generate an in-phase and a quadraturecomponents or their respective input signals.

In each vector modulator 520, 530, 540, 550, the in-phase and quadraturecomponents are multiplied with amplitude information. In FIG. 5, forexample, multiplier 524 multiplies the quadrature component of signal521 with the quadrature amplitude information I_(UY) of I_(U)(t). Inparallel, multiplier 526 multiplies the in-phase replica signal with thein-phase amplitude information sgn(I)×I_(UX) of I_(U)(t).

To generate the I_(U)(t) constant envelope constituent signals 525 and527 are summed using phase splitter 528 or alternate summing techniques.The resulting signal 529 corresponds to the IU(t) component of signalr(t).

In similar fashion as described above, vector modulators 530, 540, and550, respectively, generate the I_(L)(t), Q_(U)(t), and Q_(L)(t)components of signal r(t). I_(L)(t), Q_(U)(t), and Q_(L)(t),respectively, correspond to signals 539, 549, and 559 in FIG. 5.

Further, as described above, signals 529, 539, 549, and 559 arecharacterized by having substantially equal and constant magnitudeenvelopes. Accordingly, when signals 529, 539, 549, and 559 are inputinto corresponding power amplifiers (PA) 562, 564, 566, and 568,corresponding amplified signals 563, 565, 567, and 569 are substantiallyconstant envelope signals.

Power amplifiers 562, 564, 566, and 568 amplify each of the signals 529,539, 549, 559, respectively. In an embodiment, substantially equal poweramplification is applied to each of the signals 529, 539, 549, and 559.In an embodiment, the power amplification level of PAs 562, 564, 566,and 568 is set according to the desired power level of output signalr(t).

Still referring to FIG. 5, amplified signals 563 and 565 are summedusing summer 572 to generate an amplified version 573 of the in-phasecomponent {right arrow over (I)}(t) of signal r(t). Similarly, amplifiedsignals 567 and 569 are summed using summer 574 to generate an amplifiedversion 575 of the quadrature component Q (t) of signal r(t).

Signals 573 and 575 are summed using summer 576, as shown in FIG. 5,with the resulting signal corresponding to desired output signal r(t).

It must be noted that, in the example of FIG. 5, summers 572, 574, and576 are being used for the purpose of illustration only. Varioustechniques may be used to sum amplified signals 563, 565, 567, and 569.For example, amplified signals 563, 565, 567, and 569 may be summed allin one step to result in signal 578. In fact, according to various VPAembodiments of the present invention, it suffices that the summing isdone after amplification. Certain VPA embodiments of the presentinvention, as will be further described below, use minimally lossysumming techniques such as direct coupling via wire. Alternatively,certain VPA embodiments use conventional power combining techniques. Inother embodiments, as will be further described below, power amplifiers562, 564, 566, and 568 can be implemented as a multiple-inputsingle-output power amplifier.

Operation of the Cartesian 4-Branch VPA embodiment shall now be furtherdescribed with reference to the process flowchart of FIG. 6. The processbegins at step 610, which includes receiving the baseband representationof the desired output signal. In an embodiment, this involves receivingin-phase (I) and quadrature (Q) components of the desired output signal.In another embodiment, this involves receiving magnitude and phase ofthe desired output signal. In an embodiment of the Cartesian 4-BranchVPA embodiment, the I and Q are baseband components. In anotherembodiment, the I and Q are RF components and are down-converted tobaseband.

Step 620 includes receiving a clock signal set according to a desiredoutput signal frequency of the desired output signal. In the example ofFIG. 5, step 620 is achieved by receiving reference signal 511.

Step 630 includes processing the I component to generate first andsecond signals having the output signal frequency. The first and secondsignals have substantially constant and equal magnitude envelopes and asum equal to the I component. The first and second signals correspond tothe I_(U)(t) and I_(L)(t) constant envelope constituents describedabove. In the example of FIG. 5, step 630 is achieved by vectormodulators 520 and 530, in conjunction with their appropriate inputsignals.

Step 640 includes processing the Q component to generate third andfourth signals having the output signal frequency. The third and fourthsignals have substantially constant and equal magnitude envelopes and asum equal to the Q component. The third and fourth signals correspond tothe Q_(U)(t) and Q_(L)(t) constant envelope constituents describedabove. In the example of FIG. 5, step 630 is achieved by vectormodulators 540 and 550, in conjunction with their appropriate inputsignals.

Step 650 includes individually amplifying each of the first, second,third, and fourth signals, and summing the amplified signals to generatethe desired output signal. In an embodiment, the amplification of thefirst, second, third, and fourth signals is substantially equal andaccording to a desired power level of the desired output signal. In theexample of FIG. 5, step 650 is achieved by power amplifiers 562, 564,566, and 568 amplifying respective signals 529, 539, 549, and 559, andby summers 572, 574, and 576 summing amplified signals 563, 565, 567,and 569 to generate output signal 578.

FIG. 7A is a block diagram that illustrates an exemplary embodiment of avector power amplifier 700 implementing the process flowchart 600 ofFIG. 6. In the example of FIG. 7A, optional components are illustratedwith dashed lines. In other embodiments, additional components may beoptional.

Vector power amplifier 700 includes an in-phase (I) branch 703 and aquadrature (Q) branch 705. Each of the I and Q branches furthercomprises a first branch and a second branch.

In-phase (I) information signal 702 is received by an I Data TransferFunction module 710. In an embodiment, I information signal 702 includesa digital baseband signal. In an embodiment, I Data Transfer Functionmodule 710 samples I information signal 702 according to a sample clock706. In another embodiment, I information signal 702 includes an analogbaseband signal, which is converted to digital using ananalog-to-digital converter (ADC) (not shown in FIG. 7A) before beinginput into I Data Transfer Function module 710. In another embodiment, Iinformation signal 702 includes an analog baseband signal which input inanalog form into I Data Transfer Function module 710, which alsoincludes analog circuitry. In another embodiment, I information signal702 includes a RF signal which is down-converted to baseband beforebeing input into I Data Transfer Function module 710 using any of theabove described embodiments.

I Data Transfer Function module 710 processes I information signal 702,and determines in-phase and quadrature amplitude information of at leasttwo constant envelope constituent signals of I information signal 702.As described above with reference to FIG. 5, the in-phase and quadraturevector modulator input amplitude information corresponds tosgn(I)×I_(UX) and I_(UY), respectively. The operation of I Data TransferFunction module 710 is further described below in section 3.4.

I Data Transfer Function module 710 outputs information signals 722 and724 used to control the in-phase and quadrature amplitude components ofvector modulators 760 and 762. In an embodiment, signals 722 and 724 aredigital signals. Accordingly, each of signals 722 and 724 is fed into acorresponding digital-to-analog converter (DAC) 730 and 732,respectively. The resolution and sample rate of DACs 730 and 732 isselected to achieve the desired I component of the output signal 782.DACs 730 and 732 are controlled by DAC clock signals 723 and 725,respectively. DAC clock signals 723 and 725 may be derived from a sameclock signal or may be independent.

In another embodiment, signals 722 and 724 are analog signals, and DACs730 and 732 are not required.

In the exemplary embodiment of FIG. 7A, DACs 730 and 732 convert digitalinformation signals 722 and 724 into corresponding analog signals, andinput these analog signals into optional interpolation filters 731 and733, respectively. Interpolation filters 731 and 733, which also serveas anti-aliasing filters, shape the DACs outputs to produce the desiredoutput waveform. Interpolation filters 731 and 733 generate signals 740and 742, respectively. Signal 741 represents the inverse of signal 740.Signals 740-742 are input into vector modulators 760 and 762.

Vector modulators 760 and 762 multiply signals 740-742 withappropriately phased clock signals to generate constant envelopeconstituents of I information signal 702. The clock signals are derivedfrom a channel clock signal 708 having a rate according to a desiredoutput signal frequency. A plurality of phase splitters, such as 750 and752, for example, and phasors associated with the vector modulatormultipliers may be used to generate the appropriately phased clocksignals.

In the embodiment of FIG. 7A, for example, vector modulator 760modulates a 90° shifted channel clock signal with quadrature amplitudeinformation signal 740. In parallel, vector modulator 760 modulates anin-phase channel clock signal with in-phase amplitude information signal742. Vector modulator 760 combines the two modulated signals to generatea first modulated constant envelope constituent 761 of I informationsignal 702. Similarly, vector modulator 762 generates a second modulatedconstant envelope constituent 763 of I information signal 702, usingsignals 741 and 742. Signals 761 and 763 correspond, respectively, tothe I_(U)(t) and I_(L)(t) constant envelope components described withreference to FIG. 5.

In parallel and in similar fashion, the Q branch of vector poweramplifier 700 generates at least two constant envelope constituentsignals of quadrature (Q) information signal 704.

In the embodiment of FIG. 7A, for example, vector modulator 764generates a first constant envelope constituent 765 of Q informationsignal 704, using signals 744 and 746. Similarly, vector modulator 766generates a second constant envelope constituent 767 of Q informationsignal 704, using signals 745 and 746.

As described above with respect to FIG. 5, constituent signals 761, 763,765, and 767 have substantially equal and constant magnitude envelopes.In the exemplary embodiment of FIG. 7A, signals 761, 763, 765, and 767are, respectively, input into corresponding power amplifiers (PAs) 770,772, 774, and 776. PAs 770, 772, 774, and 776 can be linear ornon-linear power amplifiers. In an embodiment, PAs 770, 772, 774, and776 include switching power amplifiers.

Circuitry 714 and 716 (herein referred to as “autobias circuitry” forease of reference, and not limitation) and in this embodiment, controlthe bias of PAs 770, 772, 774, and 776 according to I and Q informationsignals 702 and 704. In the embodiment of FIG. 7A, autobias circuitry714 and 716 provide, respectively, bias signals 715 and 717 to PAs 770,772 and PAs 774, 776. Autobias circuitry 714 and 716 are furtherdescribed below in section 3.5. Embodiments of PAs 770, 772, 774, and776 are also discussed below in section 3.5.

In an embodiment, PAs 770, 772, 774, and 776 apply substantially equalpower amplification to respective substantially constant envelopesignals 761, 763, 765, and 767. In other embodiments, PA drivers areadditionally employed to provide additional power amplification. In theembodiment of FIG. 7A, PA drivers 794, 795, 796, and 797 are optionallyadded between respective vector modulators 760, 762, 764 766 andrespective PAs 770, 772, 774, and 776, in each branch of vector poweramplifier 700.

The outputs of PAs 770, 772, 774, and 776 are coupled together togenerate output signal 782 of vector power amplifier 700. In anembodiment, the outputs of PAs 770, 772, 774, and 776 are directlycoupled together using a wire. Direct coupling in this manner means thatthere is minimal or no resistive, inductive, or capacitive isolationbetween the outputs of PAs 770, 772, 774, and 776. In other words,outputs of PAs 770, 772, 774, and 776, are coupled together withoutintervening components. Alternatively, in an embodiment, the outputs ofPAs 770, 772, 774, and 776 are coupled together indirectly throughinductances and/or capacitances that result in low or minimal impedanceconnections, and/or connections that result in minimal isolation andminimal power loss. Alternatively, outputs of PAs 770, 772, 774, and 776are coupled using well known combining techniques, such as Wilkinson,hybrid, transformers, or known active combiners. In an embodiment, thePAs 770, 772, 774, and 776 provide integrated amplification and powercombining in a single operation. In an embodiment, one or more of thepower amplifiers and/or drivers described herein are implemented usingmultiple input, single output power amplification techniques, examplesof which are shown in FIGS. 7B, and 51A-H.

Output signal 782 includes the I and Q characteristics of I and Qinformation signals 702 and 704. Further, output signal 782 is of thesame frequency as that of its constituents, and thus is of the desiredup-converted output frequency. In embodiments of vector power amplifier700, a pull-up impedance 780 is coupled between the output of vectoramplifier 700 and a power supply. Output stage embodiments according topower amplification methods and systems of the present invention will befurther described below in section 3.5.

In other embodiments of vector power amplifier 700, process detectorsare employed to compensate for any process variations in circuitry ofthe amplifier. In the embodiment of FIG. 7A for example, processdetectors 791-793 are optionally added to monitor variations in PAdrivers 794-797 and phase splitter 750. In further embodiments,frequency compensation circuitry 799 may be employed to compensate forfrequency variations.

FIG. 7B is a block diagram that illustrates another exemplary embodimentof vector power amplifier 700. Optional components are illustrated withdashed lines, although other embodiments may have more or less optionalcomponents.

The embodiment illustrates a multiple-input single-output (MISO)implementation of the amplifier of FIG. 7A. In the embodiment of FIG.7B, constant envelope signals 761, 763, 765 and 767, output from vectormodulators 760, 762, 764, and 766, are input into MISO PAs 784 and 786.MISO PAs 784 and 786 are two-input single-output power amplifiers. In anembodiment, MISO PAs 784 and 786 include elements 770, 772, 774, 776,794-797 as shown in the embodiment of FIG. 7A or functional equivalencethereof. In another embodiment, MISO PAs 784 and 786 may include otherelements, such as optional pre-drivers and optional process detectioncircuitry. Further, MISO PAs 784 and 786 are not limited to beingtwo-input PAs as shown in FIG. 7B. In other embodiments as will bedescribed further below with reference to FIGS. 51A-H, PAs 784 and 786can have any number of inputs and outputs.

FIG. 8A is a block diagram that illustrates another exemplary embodiment800A of a vector power amplifier according to the Cartesian 4-Branch VPAmethod shown in FIG. 6. Optional components are illustrated with dashedlines, although other embodiments may have more or less optionalcomponents.

In the embodiment of FIG. 8A, a DAC 830 of sufficient resolution andsample rate replaces DACs 730, 732, 734, and 736 of the embodiment ofFIG. 7A. DAC 830's sample rate is controlled by a DAC clock signal 826.

DAC 830 receives in-phase and quadrature information signals 810 and 820from I Data Transfer Function module 710 and Q Data Transfer Functionmodule 712, respectively, as described above. In an embodiment, a inputselector 822 selects the order of signals 810 and 820 being input intoDAC 830.

DAC 830 may output a single analog signal at a time. In an embodiment, asample and hold architecture may be used to ensure proper signal timingto the four branches of the amplifier, as shown in FIG. 8A.

DAC 830 sequentially outputs analog signals 832, 834, 836, 838 to afirst set of sample-and-hold circuits 842, 844, 846, and 848. In anembodiment, DAC 830 is clocked at a sufficient rate to emulate theoperation of DACs 730, 732, 734, and 736 of the embodiment of FIG. 7A.An output selector 824 determines which of output signals 832, 834, 836,and 838 should be selected for output.

DAC830's DAC clock signal 826, output selector signal 824, inputselector 822, and sample-and-hold clocks 840A-D, and 850 are controlledby a control module that can be independent or integrated into transferfunction modules 710 and/or 712.

In an embodiment, sample-and-hold circuits (S/H) 842, 844, 846, and 848sample and hold the received analog values from DAC 830 according to aclock signals 840A-D. Sample-and-hold circuits 852, 854, 856, and 858sample and hold the analog values from sample and hold circuits 842,844, 846, and 848 respectively. In turn, sample-and-hold circuits 852,854, 856, and 858 hold the received analog values, and simultaneouslyrelease the values to vector modulators 760, 762, 764, and 766 accordingto a common clock signal 850. In another embodiment, sample-and-holdcircuits 852, 854, 856, and 858 release the values to optionalinterpolation filters 731, 733, 735, and 737 which are alsoanti-aliasing filters. In an embodiment, a common clock signal 850 isused in order to ensure that the outputs of S/H 852, 854, 856, and 858are time-aligned.

Other aspects of vector power amplifier 800A substantially correspond tothose described above with respect to vector power amplifier 700.

FIG. 8B is a block diagram that illustrates another exemplary embodiment800B of a vector power amplifier according to the Cartesian 4-Branch VPAmethod shown in FIG. 6. Optional components are illustrated with dashedlines, although other embodiments may have more or less optionalcomponents.

Embodiment 800B illustrates another single DAC implementation of thevector power amplifier. However, in contrast to the embodiment of FIG.8A, the sample and hold architecture includes a single set ofsample-and-hold (S/H) circuits. As shown in FIG. 8B, S/H 842, 844, 846,and 848 receive analog values from DAC 830, illustrated as signals 832,834, 836, and 838. Each of S/H circuits 842, 844, 846 and 848 releaseits received value according to a different clock 840A-D as shown. Thetime difference between analog samples used for to generate signals 740,741, 742, 744, 745, and 746 can be compensated for in transfer functions710 and 712. According to the embodiment of FIG. 8B, one level of S/Hcircuitry can be eliminated relative to the embodiment of FIG. 8A,thereby reducing the size and the complexity of the amplifier.

Other aspects of vector power amplifier 800B substantially correspond tothose described above with respect to vector power amplifiers 700 and800A.

FIG. 8C is a block diagram that illustrates another exemplary embodiment800C of vector power amplifier 700. Optional components are illustratedwith dashed lines, although other embodiments may have more or lessoptional components. The embodiment of FIG. 8C illustrates amultiple-input single-output (MISO) implementation of the amplifier ofFIG. 8A. In the embodiment of FIG. 8C, constant envelope signals 761,763, 765 and 767, output from vector modulators 760, 762, 764, and 766,are input into MISO PAs 860 and 862. MISO PAs 860 and 862 are two-inputsingle-output power amplifiers. In an embodiment, MISO PAs 860 and 862include elements 770, 772, 774, 776, 794-797 as shown in the embodimentof FIG. 7A or functional equivalence thereof. In another embodiment,MISO PAs 860 and 862 may include other elements, such as optionalpre-drivers and optional process detection circuitry. In anotherembodiment, MISO PAs 860 and 862 may include other elements, such aspre-drivers, not shown in the embodiment of FIG. 7A. Further, MISO PAs860 and 862 are not limited to being two-input PAs as shown in FIG. 8C.In other embodiments as will be described further below with referenceto FIGS. 51A-H, PAs 860 and 862 can have any number of inputs andoutputs.

Other aspects of vector power amplifier 800C substantially correspond tothose described above with respect to vector power amplifiers 700 and800A.

FIG. 8D is a block diagram that illustrates another exemplary embodiment800D of vector power amplifier 700. Optional components are illustratedwith dashed lines, although other embodiments may have more or lessoptional components. The embodiment of FIG. 8D illustrates amultiple-input single-output (MISO) implementation of the amplifier ofFIG. 8B. In the embodiment of FIG. 8D, constant envelope signals 761,763, 765 and 767, output from vector modulators 760, 762, 764, and 766,are input into MISO PAs 870 and 872. MISO PAs 870 and 872 are two-inputsingle-output power amplifiers. In an embodiment, MISO PAs 870 and 872include elements 770, 772, 774, 776, 794-797 as shown in the embodimentof FIG. 7A or functional equivalence thereof. In another embodiment,MISO PAs 870 and 872 may include other elements, such as optionalpre-drivers and optional process detection circuitry. In anotherembodiment, MISO PAs 870 and 872 may include other elements, such aspre-drivers, not shown in the embodiment of FIG. 7A. Further, MISO PAs870 and 872 are not limited to being two-input PAs as shown in FIG. 8D.In other embodiments as will be described further below with referenceto FIGS. 51A-H, PAs 870 and 872 can have any number of inputs andoutputs.

Other aspects of vector power amplifier 800D substantially correspond tothose described above with respect to vector power amplifiers 700 and800B.

3.2) Cartesian-Polar-Cartesian-Polar 2-Branch Vector Power Amplifier

A Cartesian-Polar-Cartesian-Polar (CPCP)-2-Branch VPA embodiment shallnow be described (The name of this embodiment is provided for ease ofreference, and is not limiting).

According to the Cartesian-Polar-Cartesian-Polar (CPCP) 2-Branch VPAmethod, a time-varying complex envelope signal is decomposed into 2substantially constant envelope constituent signals. The constituentsignals are individually amplified, and then summed to construct anamplified version of the original time-varying complex envelope signal.In addition, the phase angle of the time-varying complex envelope signalis determined and the resulting summation of the constituent signals arephase shifted by the appropriate angle.

In one embodiment of the CPCP 2-Branch VPA method, a magnitude and aphase angle of a time-varying complex envelope signal are calculatedfrom in-phase and quadrature components of a signal. Given the magnitudeinformation, two substantially constant envelope constituents arecalculated from a normalized version of the desired time-varyingenvelope signal, wherein the normalization includes implementationspecific manipulation of phase and/or amplitude. The two substantiallyconstant envelope constituents are then phase shifted by an appropriateangle related to the phase shift of the desired time-varying envelopesignal. The substantially constant envelope constituents are thenindividually amplified substantially equally, and summed to generate anamplified version of the original desired time-varying envelope signal.

FIGS. 9A and 9B conceptually illustrate the CPCP 2-Branch VPA embodimentusing a phasor signal representation. In FIG. 9A, phasor {right arrowover (R_(in))} represents a time-varying complex envelope input signalr(t). At any instant of time, {right arrow over (R_(in))} reflects amagnitude and a phase shift angle of signal r(t). In the example shownin FIG. 9A, {right arrow over (R_(in))} is characterized by a magnitudeR and a phase shift angle θ. As described above, the phase shift angleis measured relative to a reference signal.

Referring to FIG. 9A, {right arrow over (R′)} represents the relativeamplitude component of {right arrow over (R)}_(in) generated by {rightarrow over (U)}′ and {right arrow over (L)}′.

Still referring to FIG. 9A, it is noted that, at any time instant,{right arrow over (R′)} can be obtained by the sum of an upper phasor{right arrow over (U′)} and a lower phasor {right arrow over (L′)}.Further, {right arrow over (U′)} and {right arrow over (L′)} can bemaintained to have substantially constant magnitude. The phasors, {rightarrow over (U′)} and {right arrow over (L′)}, accordingly, represent twosubstantially constant envelope signals. r′(t) can thus be obtained, atany time instant, by the sum of two substantially constant envelopesignals that correspond to phasors {right arrow over (U′)} and {rightarrow over (L′)}.

The phase shifts of phasors {right arrow over (U′)} and {right arrowover (L′)} relative to {right arrow over (R′)} are set according to thedesired magnitude R of {right arrow over (R′)}. In the simplest case,when upper and lower phasors {right arrow over (U′)} and {right arrowover (L′)} are selected to have equal magnitude, upper and lower phasors{right arrow over (U′)} and {right arrow over (L′)} are substantiallysymmetrically shifted in phase relative to {right arrow over (R′)}. Thisis illustrated in the example of FIG. 9A. It is noted that terms andphrases indicating or suggesting orientation, such as but not limited to“upper and lower” are used herein for ease of reference and are notfunctionally or structurally limiting.

It can be verified that, for the case illustrated in FIG. 9A, the phaseshift of {right arrow over (U′)} and {right arrow over (L′)} relative to{right arrow over (R′)}, illustrated as angle

$\frac{\phi}{2}$in FIG. 9A, is related to the magnitude of {right arrow over (R′)} asfollows:

$\begin{matrix}{\frac{\phi}{2} = {\cot^{- 1}\left( \frac{R}{2\sqrt{1 - \frac{R^{2}}{4}}} \right)}} & (7)\end{matrix}$where R represents a normalized magnitude of phasor {right arrow over(R′)}.

Equation (7) can further be reduced to

$\begin{matrix}{\frac{\phi}{2} = {\cos^{- 1}\left( \frac{R}{2} \right)}} & (7.10)\end{matrix}$where R represents a normalized magnitude of phasor {right arrow over(R′)}.

Alternatively, any substantially equivalent mathematical equations orother substantially equivalent mathematical techniques such as look uptables can be used.

It follows from the above discussion that, in phasor representation, anyphasor {right arrow over (R′)} of variable magnitude and phase can beconstructed by the sum of two constant magnitude phasor components:{right arrow over (R′)}={right arrow over (U′)}+{right arrow over (L′)}|{right arrow over (U)}|=|{right arrow over (L)}|=A=constant  (8)

Correspondingly, in the time domain, a time-varying envelope sinusoidalsignal r′(t)=R(t)×cos(ωt) is constructed by the sum of two constantenvelope signals as follows:

$\begin{matrix}{{{r^{\prime}(t)} = {{U^{\prime}(t)} + {L^{\prime}(t)}}}{{{U^{\prime}(t)} = {A \times {\cos\left( {{\omega\; t} + \frac{\phi}{2}} \right)}}};}{{{L^{\prime}(t)} = {A \times \cos\left( {{\omega\; t} - \frac{\phi}{2}} \right)}};}} & (9)\end{matrix}$where A is a constant and

$\frac{\phi}{2}$is as shown in equation (7).

From FIG. 9A, it can be further verified that equations (9) can berewritten as:r′(t)=U′(t)+L′(t);U′(t)=C cos(ωt)+α sin(ωt);L′(t)=C cos(ωt)−β sin(ωt);  (10)where C denotes the real part component of phasors {right arrow over(U′)} and {right arrow over (L′)} and is equal to

$A \times {{\cos\left( \frac{\phi}{2} \right)}.}$Note that C is a common component of {right arrow over (U′)} and {rightarrow over (L′)}. α and β denote the imaginary part components ofphasors {right arrow over (U′)} and {right arrow over (L′)},respectively.

$\alpha = {\beta = {A \times {{\sin\left( \frac{\phi}{2} \right)}.}}}$Accordingly, from equations (12),

${r^{\prime}(t)} = {{2\; C \times {\cos\left( {\omega\; t} \right)}} = {2A \times {\cos\left( \frac{\phi}{2} \right)} \times {{\cos\left( {\omega\; t} \right)}.}}}$As understood by a person skilled in the art based on the teachingsherein, other equivalent and/or simplified representations of the aboverepresentations of the quantities A, B, and C may also be used,including look up tables, for example.

Note that {right arrow over (R_(in))} is shifted by θ degrees relativeto {right arrow over (R′)}. Accordingly, using equations (8), it can bededuced that:{right arrow over (R _(in))}={right arrow over (R′)}e ^(jθ)=({rightarrow over (U′)}+{right arrow over (L′)})e ^(jθ) ={right arrow over(U′)}e ^(jθ) +{right arrow over (L′)}e ^(jθ)  (11)

Equations (11) imply that a representation of {right arrow over(R_(in))} can be obtained by summing phasors {right arrow over (U′)} and{right arrow over (L′)}, described above, shifted by θ degrees. Further,an amplified output version, {right arrow over (R_(out))}, of {rightarrow over (R_(in))} can be obtained by separately amplifyingsubstantially equally each of the θ degrees shifted versions of phasors{right arrow over (U′)} and {right arrow over (L′)}, and summing them.FIG. 9B illustrates this concept. In FIG. 9B, phasors {right arrow over(U)} and {right arrow over (L)} represent θ degrees shifted andamplified versions of phasors {right arrow over (U′)} and {right arrowover (L′)}. Note that, since {right arrow over (U′)} and {right arrowover (L′)} are constant magnitude phasors, {right arrow over (U)} and{right arrow over (L)} are also constant magnitude phasors. Phasors{right arrow over (U)} and {right arrow over (L)} sum, as shown FIG. 9B,to phasor {right arrow over (R_(out))}, which is a power amplifiedversion of input signal {right arrow over (R_(in))}.

Equivalently, in the time domain, it can be shown that:r _(out)(t)=U(t)+L(t);U(t)=K[C cos(ωt+θ)+α sin(ωt+θ)];L(t)=K[C cos(ωt+θ)−β sin(ωt+θ)].  (12)where r_(out)(t) corresponds to the time domain signal represented byphasor {right arrow over (R_(out))}, U(t) and L(t) correspond to thetime domain signals represents by phasors {right arrow over (U)} and{right arrow over (L)}, and K is the power amplification factor.

A person skilled in the art will appreciate that, whereas the timedomain representations in equations (9) and (10) have been provided forthe case of a sinusoidal waveform, equivalent representations can bedeveloped for non-sinusoidal waveforms using appropriate basisfunctions.

FIG. 10 is a block diagram that conceptually illustrates an exemplaryembodiment 1000 of the CPCP 2-Branch VPA embodiment. An output signalr(t) of desired power level and frequency characteristics is generatedfrom in-phase and quadrature components according to the CPCP 2-BranchVPA embodiment.

In the example of FIG. 10, a clock signal 1010 represents a referencesignal for generating output signal r(t). Clock signal 1010 is of thesame frequency as that of desired output signal r(t).

Referring to FIG. 10, an Iclk_phase signal 1012 and a Qclk_phase signal1014 represent amplitude analog values that are multiplied by thein-phase and quadrature components of Clk signal 1010 and are calculatedfrom the baseband I and Q signals.

Still referring to FIG. 10, clock signal 1010 is multiplied withIclk_phase signal 1012. In parallel, a 90° degrees shifted version ofclock signal 1010 is multiplied with Qclk_phase signal 1014. The twomultiplied signals are combined to generate Rclk signal 1016. Rclksignal 1016 is of the same frequency as clock signal 1010. Further, Rclksignal 1016 is characterized by a phase shift angle according to theratio of Q(t) and I(t). The magnitude of Rclk signal 1016 is such thatR²clk=I²clk_phase+Q²clk_phase. Accordingly, Rclk signal 1016 representsa substantially constant envelope signal having the phasecharacteristics of the desired output signal r(t).

Still referring to FIG. 10, Rclk signal 1016 is input, in parallel, intotwo vector modulators 1060 and 1062. Vector modulators 1060 and 1062generate the U(t) and L(t) substantially constant envelope constituents,respectively, of the desired output signal r(t) as described in (12). Invector modulator 1060, an in-phase Rclk signal 1020, multiplied withCommon signal 1028, is combined with a 90° degree shifted version 1018of Rclk signal, multiplied with first signal 1026. In parallel, invector modulator 1062, an in-phase Rclk signal 1022, multiplied withCommon signal 1028, is combined with a 90° degrees shifted version 1024of Rclk signal, multiplied with second signal 1030. Common signal 1028,first signal 1026, and second signal 1030 correspond, respectively, tothe real part C and the imaginary parts α and β described in equation(12).

Output signals 1040 and 1042 of respective vector modulators 1060 and1062 correspond, respectively, to the U(t) and L(t) constant envelopeconstituents of input signal r(t).

As described above, signals 1040 and 1042 are characterized by havingsubstantially equal and constant magnitude envelopes. Accordingly, whensignals 1040 and 1042 are input into corresponding power amplifiers (PA)1044 and 1046, corresponding amplified signals 1048 and 1050 aresubstantially constant envelope signals.

Power amplifiers 1044 and 1046 apply substantially equal poweramplification to signals 1040 and 1042, respectively. In an embodiment,the power amplification level of PAs 1044 and 1046 is set according tothe desired power level of output signal r(t). Further, amplifiedsignals 1048 and 1050 are in-phase relative to each other. Accordingly,when summed together, as shown in FIG. 10, resulting signal 1052corresponds to the desired output signal r(t).

FIG. 10A is another exemplary embodiment 1000A of the CPCP 2-Branch VPAembodiment. Embodiment 1000A represents a Multiple Input Single Output(MISO) implementation of embodiment 1000 of FIG. 10.

In embodiment 1000A, constant envelope signals 1040 and 1042, outputfrom vector modulators 1060 and 1062, are input into MISO PA 1054. MISOPA 1054 is a two-input single-output power amplifier. In an embodiment,MISO PA 1054 may include various elements, such as pre-drivers, drivers,power amplifiers, and process detectors (not shown in FIG. 10A), forexample. Further, MISO PA 1054 is not limited to being a two-input PA asshown in FIG. 10A. In other embodiments, as will be described furtherbelow with reference to FIGS. 51A-H, PA 1054 can have any number ofinputs.

Operation of the CPCP 2-Branch VPA embodiment is depicted in the processflowchart 1100 of FIG. 11.

The process begins at step 1110, which includes receiving a basebandrepresentation of the desired output signal. In an embodiment, thisinvolves receiving in-phase (I) and quadrature (Q) components of thedesired output signal. In another embodiment, this involves receivingmagnitude and phase of the desired output signal.

Step 1120 includes receiving a clock signal set according to a desiredoutput signal frequency of the desired output signal. In the example ofFIG. 10, step 1120 is achieved by receiving clock signal 1010.

Step 1130 includes processing the clock signal to generate a normalizedclock signal having a phase shift angle according to the received I andQ components. In an embodiment, the normalized clock signal is aconstant envelope signal having a phase shift angle according to a ratioof the I and Q components. The phase shift angle of the normalized clockis relative to the original clock signal. In the example of FIG. 10,step 1130 is achieved by multiplying clock signal 1010's in-phase andquadrature components with Iclk_phase 1012 and Qclk_phase 1014 signals,and then summing the multiplied signal to generate Rclk signal 1016.

Step 1140 includes the processing of the I and Q components to generatethe amplitude information required to produce first and secondsubstantially constant envelope constituent signals.

Step 1150 includes processing the amplitude information of step 1140 andthe normalized clock signal Rclk to generate the first and secondconstant envelope constituents of the desired output signal. In anembodiment, step 1150 involves phase shifting the first and secondconstant envelope constituents of the desired output signal by the phaseshift angle of the normalized clock signal. In the example of FIG. 10,step 1150 is achieved by vector modulators 1060 and 1062 modulating Rclksignal 1016 with first signal 1026, second signal 1030, and commonsignal 1028 to generate signals 1040 and 1042.

Step 1160 includes individually amplifying the first and second constantenvelope constituents, and summing the amplified signals to generate thedesired output signal. In an embodiment, the amplification of the firstand second constant envelope constituents is substantially equal andaccording to a desired power level of the desired output signal. In theexample of FIG. 10, step 1160 is achieved by PAs 1044 and 1046amplifying signals 1040 and 1042 to generate amplified signals 1048 and1050.

FIG. 12 is a block diagram that illustrates an exemplary embodiment of avector power amplifier 1200 implementing the process flowchart 1100.Optional components are illustrated with dashed lines, although in otherembodiments more or less components may be optional.

Referring to FIG. 12, in-phase (I) and quadrature (Q) information signal1210 is received by an I and Q Data Transfer Function module 1216. In anembodiment, I and Q Data Transfer Function 1216 samples signal 1210according to a sample clock 1212. I and Q information signal 1210includes baseband I and Q information of a desired output signal r(t).

In an embodiment, I and Q Data Transfer Function module 1216 processesinformation signal 1210 to generate information signals 1220, 1222,1224, and 1226. The operation of I and Q Data Transfer Function module1216 is further described below in section 3.4.

Referring to FIG. 12, information signal 1220 includes quadratureamplitude information of first and second constant envelope constituentsof a baseband version of desired output signal r(t). With reference toFIG. 9A, for example, information signal 1220 includes the α and βquadrature components. Referring again to FIG. 12, information signal1226 includes in-phase amplitude information of the first and secondconstant envelope constituents of the baseband version of signal r(t).With reference to FIG. 9A, for example, information signal 1226 includesthe common C in-phase component.

Still referring to FIG. 12, information signals 1222 and 1224 includenormalized in-phase Iclk_phase and quadrature Qclk_phase signals,respectively. Iclk_phase and Qclk_phase are normalized versions of the Iand Q information signals included in signal 1210. In an embodiment,Iclk_phase and Qclk_phase are normalized such that that(I²clk_phase+Q²clk_phase=constant). It is noted that the phase of signal1250 corresponds to the phase of the desired output signal and iscreated from Iclk_phase and Qclk_phase. Referring to FIG. 9B, Iclk_phaseand Qclk_phase are related to I and Q as follows:

$\begin{matrix}{\theta = {{\tan^{- 1}\left( \frac{Q}{I} \right)} = {\tan^{- 1}\left( \frac{Q_{clk\_ phase}}{I_{clk\_ phase}} \right)}}} & (12.1)\end{matrix}$where θ represents the phase of the desired output signal, represented bphasor {right arrow over (R_(out))} in FIG. 9B. The sign information ofthe baseband I and Q information must be taken into account to calculateθ for all four quadrants.

In the exemplary embodiment of FIG. 12, information signals 1220, 1222,1224, and 1226 are digital signals. Accordingly, each of signals 1220,1222, 1224, and 1226 is fed into a corresponding digital-to-analogconverter (DAC) 1230, 1232, 1234, and 1236. The resolution and samplerate of DACs 1230, 1232, 1234, and 1236 is selected according tospecific signaling schemes. DACs 1230, 1232, 1234, and 1236 arecontrolled by DAC clock signals 1221, 1223, 1225, and 1227,respectively. DAC clock signals 1221, 1223, 1225, and 1227 may bederived from a same clock signal or may be independent.

In other embodiments, information signals 1220, 1222, 1224, and 1226 aregenerated in analog format and no DACs are required.

Referring to FIG. 12, DACs 1230, 1232, 1234, and 1236 convert digitalinformation signals 1220, 1222, 1224, and 1226 into corresponding analogsignals, and input these analog signal into optional interpolationfilters 1231, 1233, 1235, and 1237, respectively. Interpolation filters1231, 1233, 1235, and 1237, which also serve as anti-aliasing filters,shape the DACs output signals to produce the desired output waveform.Interpolation filters 1231, 1233, 1235, and 1237 generate signals 1240,1244, 1246, and 1248, respectively. Signal 1242 represents the inverseof signal 1240.

Still referring to FIG. 12, signals 1244 and 1246, which includeIclk_phase and Qclk_phase information, are input into a vector modulator1238. Vector modulator 1238 multiplies signal 1244 with a channel clocksignal 1214. Channel clock signal 1214 is selected according to adesired output signal frequency. In parallel, vector modulator 1238multiplies signal 1246 with a 90° shifted version of channel clocksignal 1214. In other words, vector modulator 1238 generates an in-phasecomponent having amplitude of Iclk_phase and a quadrature componenthaving amplitude of Qclk_phase.

Vector modulator 1238 combines the two modulated signals to generateRclk signal 1250. Rclk signal 1250 is a substantially constant envelopesignal having the desired output frequency and a phase shift angleaccording to the I and Q data included in signal 1210.

Still referring to FIG. 12, signals 1240, 1242, and 1248 include the U,L, and Common C amplitude components, respectively, of the complexenvelope of signal r(t). Signals 1240, 1242, and 1248 along with Rclksignal 1250 are input into vector modulators 1260 and 1262.

Vector modulator 1260 combines signal 1240, multiplied with a 90°shifted version of Rclk signal 1250, and signal 1248, multiplied with a0° shifted version of Rclk signal 1250, to generate output signal 1264.In parallel, vector modulator 1262 combines signal 1242, multiplied witha 90° shifted version of Rclk signal 1250, and signal 1248, modulatedwith a 0° shifted version of Rclk signal 1250, to generate output signal1266.

Output signals 1264 and 1266 represent substantially constant envelopesignals. Further, phase shifts of output signals 1264 and 1266 relativeto Rclk signal 1250 are determined by the angle relationships associatedwith the ratios α/C and β/C, respectively. In an embodiment, α=−β andtherefore output signals 1264 and 1266 are symmetrically phased relativeto Rclk signal 1250. With reference to FIG. 9B, for example, outputsignals 1264 and 1266 correspond, respectively, to the {right arrow over(U)} and {right arrow over (L)} constant magnitude phasors.

A sum of output signals 1264 and 1266 results in achannel-clock-modulated signal having the I and Q characteristics ofbaseband signal r(t). To achieve a desired power level at the output ofvector power amplifier 1200, however, signals 1264 and 1266 areamplified to generate an amplified output signal. In the embodiment ofFIG. 12, signals 1264 and 1266 are, respectively, input into poweramplifiers (PAs) 1270 and 1272 and amplified. In an embodiment, PAs 1270and 1272 include switching power amplifiers. Autobias circuitry 1218controls the bias of PAs 1270 and 1272 as further described below insection 3.5.2. In the embodiment of FIG. 12, for example, autobiascircuitry 1218 provides a bias voltage 1228 to PAs 1270 and 1272.

In an embodiment, PAs 1270 and 1272 apply substantially equal poweramplification to respective constant envelope signals 1264-1266. In anembodiment, the power amplification is set according to the desiredoutput power level. In other embodiments of vector power amplifier 1200,PA drivers and/or pre-drivers are additionally employed to provideadditional power amplification capability to the amplifier. In theembodiment of FIG. 12, for example, PA drivers 1284 and 1286 areoptionally added, respectively, between vector modulators 1260 and 1262and subsequent PAs 1270 and 1272.

Respective output signals 1274 and 1276 of PAs 1270 and 1272 aresubstantially constant envelope signals. Further, when output signals1274 and 1276 are summed, the resulting signal has minimal non-lineardistortion. In the embodiment of FIG. 12, output signals 1274 and 1276are coupled together to generate output signal 1280 of vector poweramplifier 1200. In an embodiment, no isolation is used in coupling theoutputs of PAs 1270 and 1272. Accordingly, minimal power loss isincurred by the coupling. In an embodiment, the outputs of PAs 1270 and1272 are directly coupled together using a wire. Direct coupling in thismanner means that there is minimal or no resistive, inductive, orcapacitive isolation between the outputs of PAs 1270 and 1272. In otherwords, outputs of PAs 1270 and 1272 are coupled together withoutintervening components. Alternatively, in an embodiment, the outputs ofPAs 1270 and 1272 are coupled together indirectly through inductancesand/or capacitances that result in low or minimal impedance connections,and/or connections that result in minimal isolation and minimal powerloss. Alternatively, outputs of PAs 1270 and 1272 are coupled using wellknown combining techniques, such as Wilkinson, hybrid combiners,transformers, or known active combiners. In an embodiment, the PAs 1270and 1272 provide integrated amplification and power combining in asingle operation. In an embodiment, one or more of the power amplifiersand/or drivers described herein are implemented using multiple input,single output power amplification techniques, examples of which areshown in FIGS. 12A, 12B, and 51A-H.

Output signal 1280 represents a signal having the I and Qcharacteristics of baseband signal r(t) and the desired output powerlevel and frequency. In embodiments of vector power amplifier 1200, apull-up impedance 1288 is coupled between the output of vector poweramplifier 1200 and a power supply. In other embodiments, an impedancematching network 1290 is coupled at the output of vector power amplifier1200. Output stage embodiments according to power amplification methodsand systems of the present invention will be further described below insection 3.5.

In other embodiments of vector power amplifier 1200, process detectorsare employed to compensate for any process variations in circuitry ofthe amplifier. In the exemplary embodiment of FIG. 12, for example,process detector 1282 is optionally added to monitor variations in PAdrivers 1284 and 1286.

FIG. 12A is a block diagram that illustrates another exemplaryembodiment of a vector power amplifier 1200A implementing the processflowchart 1100. Optional components are illustrated with dashed lines,although in other embodiments more or less components may be optional.

Embodiment 1200A illustrates a multiple-input single-output (MISO)implementation of embodiment 1200. In embodiment 1200A, constantenvelope signals 1261 and 1263, output from vector modulators 1260 and1262, are input into MISO PA 1292. MISO PA 1292 is a two-inputsingle-output power amplifier. In an embodiment, MISO PA 1292 includeselements 1270, 1272, 1282, 1284, and 1286 as shown in the embodiment ofFIG. 12. In another embodiment, MISO PA 1292 may include other elements,such as pre-drivers, not shown in the embodiment of FIG. 12. Further,MISO PA 1292 is not limited to being a two-input PA as shown in FIG.12A. In other embodiments as will be described further below withreference to FIGS. 51A-H, PA 1292 can have any number of inputs andoutputs.

Still referring to FIG. 12A, embodiment 1200A illustrates oneimplementation for delivering autobias signals to MISO PA 1292. In theembodiment of FIG. 12A, Autobias signal 1228 generated by Autobiascircuitry 1218, has one or more signals derived from it to biasdifferent stages of MISO PA 1292. As shown in the example of FIG. 12A,three bias control signals Bias A, Bias B, and Bias C are derived fromAutobias signal 1228, and then input at different stages of MISO PA1292. For example, Bias C may be the bias signal to the pre-driver stageof MISO PA 1292. Similarly, Bias B and Bias A may be the bias signals tothe driver and PA stages of MISO PA 1292.

In another implementation, shown in embodiment 1200B of FIG. 12 B,Autobias circuitry 1218 generates separate Autobias signals 1295, 1296,and 1295, corresponding to Bias A, Bias B, and Bias C, respectively.Signals 1295, 1296, and 1297 may or may not be generated separatelywithin Autobias circuitry 1218, but are output separately as shown.Further, signals 1295, 1296, and 1297 may or may not be related asdetermined by the biasing of the different stages of MISO PA 1294.

Other aspects of vector power amplifiers 1200A and 1200B substantiallycorrespond to those described above with respect to vector poweramplifier 1200.

FIG. 13 is a block diagram that illustrates another exemplary embodiment1300 of a vector power amplifier according to the CPCP 2-Branch VPAembodiment. Optional components are illustrated with dashed lines,although in other embodiments more or less components may be optional.

In the exemplary embodiment of FIG. 13, a DAC of sufficient resolutionand sample rate 1320 replaces DACs 1230, 1232, 1234 and 1236 of theembodiment of FIG. 12. DAC 1320 is controlled by a DAC clock 1324.

DAC 1320 receives information signal 1310 from I and Q Data TransferFunction module 1216. Information signal 1310 includes identicalinformation content to signals 1220, 1222, 1224 and 1226 in theembodiment of FIG. 12.

DAC 1320 may output a single analog signal at a time. Accordingly, asample-and-hold architecture may be used as shown in FIG. 13.

DAC 1320 sequentially outputs analog signals 1332, 1334, 1336, 1336 to afirst set of sample-and-hold circuits 1342, 1344, 1346, and 1348. In anembodiment, DAC 1230 is clocked at a sufficient rate to replace DACs1230, 1232, 1234, and 1236 of the embodiment of FIG. 12. An outputselector 1322 determines which of output signals 1332, 1334, 1336, and1338 should be selected for output.

DAC1320's DAC clock signal 1324, output selector signal 1322, andsample-and-hold clocks 1340A-D and 1350 are controlled by a controlmodule that can be independent or integrated into transfer functionmodule 1216.

In an embodiment, sample-and-hold circuits (S/H) 1342, 1344, 1346, and1348 hold the received analog values and, according to a clock signal1340A-D, release the values to a second set of sample-and-hold circuits1352, 1354, 1356, and 1358. For example, S/H 1342 release its value toS/H 1352 according to a received clock signal 1340A. In turn,sample-and-hold circuits 1352, 1354, 1356, and 1358 hold the receivedanalog values, and simultaneously release the values to interpolationfilters 1231, 1233, 1235, and 1237 according to a common clock signal1350. A common clock signal 1350 is used in order to ensure that theoutputs of S/H 1352, 1354, 1356, and 1358 are time-aligned.

In another embodiment, a single layer of S/H circuitry that includes S/H1342, 1344, 1346, and 1348 can be employed. Accordingly, S/H circuits1342, 1344, 1346, and 1348 receive analog values from DAC 1320, and eachreleases its received value according to a clock independent of theothers. For example, S/H 1342 is controlled by clock 1340A, which maynot be synchronized with clock 1340B that controls S/H 1344. To ensurethat outputs of S/H circuits 1342, 1344, 1346, and 1348 aretime-aligned, delays between clocks 1340A-D are pre-compensated for inprior stages of the amplifier. For example, DAC 1320 outputs signal1332, 1334, 1336, and 1338 with appropriately selected delays to S/Hcircuits 1342, 1344, 1346, and 1348 in order to compensate for the timedifferences between clocks 1340A-D.

Other aspects of vector power amplifier 1300 are substantiallyequivalent to those described above with respect to vector poweramplifier 1200.

FIG. 13A is a block diagram that illustrates another exemplaryembodiment 1300A of a vector power amplifier according to the CPCP2-Branch VPA embodiment. Optional components are illustrated with dashedlines, although in other embodiments more or less components may beoptional. Embodiment 1300A is a MISO implementation of embodiment 1300of FIG. 13.

In the embodiment of FIG. 13A, constant envelope signals 1261 and 1263output from vector modulators 1260 and 1262 are input into MISO PA 1360.MISO PA 1360 is a two-input single-output power amplifier. In anembodiment, MISO PA 1360 includes elements 1270, 1272, 1282, 1284, and1286 as shown in the embodiment of FIG. 13. In another embodiment, MISOPA 1360 may include other elements, such as pre-drivers, not shown inthe embodiment of FIG. 13, or functional equivalents thereof. Further,MISO PA 1360 is not limited to being a two-input PA as shown in FIG.13A. In other embodiments as will be described further below withreference to FIGS. 51A-H, PA 1360 can have any number of inputs.

The embodiment of FIG. 13A further illustrates two different sample andhold architectures with a single or two levels of S/H circuitry asshown. The two implementations have been described above with respect toFIG. 13.

Embodiment 1300A also illustrates optional bias control circuitry 1218and associated bias control signal 1325, 1326, and 1327. Signals 1325,1326, and 1327 may be used to bias different stages of MISO PA 1360 incertain embodiments.

Other aspects of vector power amplifier 1300A are equivalent to thosedescribed above with respect to vector power amplifiers 1200 and 1300.

3.3) Direct Cartesian 2-Branch Vector Power Amplifier

A Direct Cartesian 2-Branch VPA embodiment shall now be described. Thisname is used herein for reference purposes, and is not functionally orstructurally limiting.

According to the Direct Cartesian 2-Branch VPA embodiment, atime-varying envelope signal is decomposed into two constant envelopeconstituent signals. The constituent signals are individually amplifiedequally or substantially equally, and then summed to construct anamplified version of the original time-varying envelope signal.

In one embodiment of the Direct Cartesian 2-Branch VPA embodiment, amagnitude and a phase angle of a time-varying envelope signal arecalculated from in-phase and quadrature components of an input signal.Using the magnitude and phase information, in-phase and quadratureamplitude components are calculated for two constant envelopeconstituents of the time-varying envelope signal. The two constantenvelope constituents are then generated, amplified equally orsubstantially equally, and summed to generate an amplified version ofthe original time-varying envelope signal R_(in).

The concept of the Direct Cartesian 2-Branch VPA will now be describedwith reference to FIGS. 9A and 14.

As described and verified above with respect to FIG. 9A, the phasor{right arrow over (R′)} can be obtained by the sum of an upper phasor{right arrow over (U′)} and a lower phasor {right arrow over (L′)}appropriately phased to produce {right arrow over (R′)}. {right arrowover (R′)} is calculated to be proportional to the magnitude R_(in).Further, {right arrow over (U′)} and {right arrow over (L′)} can bemaintained to have substantially constant magnitude. In the time domain,{right arrow over (U′)} and {right arrow over (L′)} represent twosubstantially constant envelope signals. The time domain equivalentr′(t) of {right arrow over (R′)} can thus be obtained, at any timeinstant, by the sum of two substantially constant envelope signals.

For the case illustrated in FIG. 9A, the phase shift of {right arrowover (U′)} and {right arrow over (L′)} relative to {right arrow over(R′)}, illustrated as angle

$\frac{\phi}{2}$in FIG. 9A, is related to the magnitude of {right arrow over (R′)} asfollows:

$\begin{matrix}{\frac{\phi}{2} = {\cot^{- 1}\left( \frac{R}{2\sqrt{1 - \frac{R^{2}}{4}}} \right)}} & (13)\end{matrix}$where R represents the normalized magnitude of phasor {right arrow over(R′)}.

In the time domain, it was shown that a time-varying envelope signal,r′(t)=R(t)cos(ωt) for example, can be constructed by the sum of twoconstant envelope signals as follows:r′(t)=U′(t)+L′(t);U′(t)=C×cos(ωt)+α×sin(ωt);L′(t)=C×cos(ωt)−β×sin(ωt).  (14)where C denotes the in-phase amplitude component of phasors {right arrowover (U′)} and {right arrow over (L′)} and is equal or substantiallyequal to

$A \times {\cos\left( \frac{\phi}{2} \right)}$(A being a constant). α and β denote the quadrature amplitude componentsof phasors {right arrow over (U′)} and {right arrow over (L′)},respectively.

$\alpha = {\beta = {A \times {{\sin\left( \frac{\phi}{2} \right)}.}}}$Note that equations (14) can be modified for non-sinusoidal signals bychanging the basis function from sinusoidal to the desired function.

FIG. 14 illustrates phasor {right arrow over (R)} and its two constantmagnitude constituent phasors {right arrow over (U)} and {right arrowover (L)}. {right arrow over (R)} is shifted by θ degrees relative to{right arrow over (R′)} in FIG. 9A. Accordingly, it can be verifiedthat:{right arrow over (R)}={right arrow over (R′)}×e ^(jθ)=({right arrowover (U′)}+{right arrow over (L′)})×e ^(jθ) ={right arrow over(U)}+{right arrow over (L)};{right arrow over (U)}={right arrow over (U′)}×e ^(jθ);{right arrow over (L)}={right arrow over (L′)}×e ^(jθ).  (15)

From equations (15), it can be further shown that:{right arrow over (U)}={right arrow over (U′)}×e ^(jθ)=(C+jα)×e ^(jθ);

{right arrow over (U)}=(C+jα)(cos θ+j sin θ)=(C cos θ−α sin θ)+j(C sinθ+α cos θ).  (16)

Similarly, it can be shown that:{right arrow over (L)}={right arrow over (L′)}×e ^(jθ)=(C+jβ)×e ^(jθ);

{right arrow over (L)}=(C+jβ)(cos θ+j sin θ)=(C cos θ−β sin θ)+j(C sinθ+β cos θ).  (17)

Equations (16) and (17) can be re-written as:{right arrow over (U)}=(C cos θ−α sin θ)+j(C sin θ+α cos θ)=U _(x) +jU_(y);{right arrow over (L)}=(C cos θ−β sin θ)+j(C sin θ+β cos θ)=L _(x) +jL_(y).  (18)

Equivalently, in the time domain:U(t)=U _(x)φ₁(t)+U _(y)φ₂(t);L(t)=L _(x)φ₁(t)+L _(y)φ₂(t);  (19)

where φ₁(t) and φ₂(t) represent an appropriately selected orthogonalbasis functions.

From equations (18) and (19), it is noted that it is sufficient tocalculate the values of α, β, C and sin(Θ) and cos(Θ) in order todetermine the two constant envelope constituents of a time-varyingenvelope signal r(t). Further, α, β and C can be entirely determinedfrom magnitude and phase information, equivalently I and Q components,of signal r(t).

FIG. 15 is a block diagram that conceptually illustrates an exemplaryembodiment 1500 of the Direct Cartesian 2-Branch VPA embodiment. Anoutput signal r(t) of desired power level and frequency characteristicsis generated from in-phase and quadrature components according to theDirect Cartesian 2-Branch VPA embodiment.

In the example of FIG. 15, a clock signal 1510 represents a referencesignal for generating output signal r(t). Clock signal 1510 is of thesame frequency as that of desired output signal r(t).

Referring to FIG. 15, exemplary embodiment 1500 includes a first branch1572 and a second branch 1574. The first branch 1572 includes a vectormodulator 1520 and a power amplifier (PA) 1550. Similarly, the secondbranch 1574 includes a vector modulator 1530 and a power amplifier (PA)1560.

Still referring to FIG. 15, clock signal 1510 is input, in parallel,into vector modulators 1520 and 1530. In vector modulator 1520, anin-phase version 1522 of clock signal 1510, multiplied with U_(x) signal1526, is summed with a 90° degrees shifted version 1524 of clock signal1510, multiplied with U_(y) signal 1528. In parallel, in vectormodulator 1530, an in-phase version 1532 of clock signal 1510,multiplied with Lx signal 1536, is summed with a 90° degrees shiftedversion 1534 of clock signal 1510, multiplied with Ly signal 1538. U_(x)signal 1526 and U_(y) signal 1528 correspond, respectively, to thein-phase and quadrature amplitude components of the U(t) constantenvelope constituent of signal r(t) provided in equation (19).Similarly, L_(x) signal 1536, and L_(y) signal 1538 correspond,respectively, to the in-phase and quadrature amplitude components of theL(t) constant envelope constituent of signal r(t) provided in equation(19).

Accordingly, respective output signals 1540 and 1542 of vectormodulators 1520 and 1530 correspond, respectively, to the U(t) and L(t)constant envelope constituents of signal r(t) as described above inequations (19). As described above, signals 1540 and 1542 arecharacterized by having equal and constant or substantially equal andconstant magnitude envelopes.

Referring to FIG. 15, to generate the desired power level of outputsignal r(t), signals 1540 and 1542 are input into corresponding poweramplifiers 1550 and 1560.

In an embodiment, power amplifiers 1550 and 1560 apply equal orsubstantially equal power amplification to signals 1540 and 1542,respectively. In an embodiment, the power amplification level of PAs1550 and 1560 is set according to the desired power level of outputsignal r(t).

Amplified output signals 1562 and 1564 are substantially constantenvelope signals. Accordingly, when summed together, as shown in FIG.15, resulting signal 1570 corresponds to the desired output signal r(t).

FIG. 15A is another exemplary embodiment 1500A of the Direct Cartesian2-Branch VPA embodiment. Embodiment 1500A represents a Multiple InputSignal Output (MISO) implementation of embodiment 1500 of FIG. 15.

In embodiment 1500A, constant envelope signals 1540 and 1542, outputfrom vector modulators 1520 and 1530, are input into MISO PA 1580. MISOPA 1580 is a two-input single-output power amplifier. In an embodiment,MISO PA 1580 may include various elements, such as pre-drivers, drivers,power amplifiers, and process detectors (not shown in FIG. 15A), forexample. Further, MISO PA 1580 is not limited to being a two-input PA asshown in FIG. 15A. In other embodiments, as will be described furtherbelow with reference to FIGS. 51A-H, PA 1580 can have any number ofinputs.

Operation of the Direct Cartesian 2-Branch VPA embodiment is depicted inthe process flowchart 1600 of FIG. 16. The process begins at step 1610,which includes receiving a baseband representation of a desired outputsignal. In an embodiment, the baseband representation includes I and Qcomponents. In another embodiment, the I and Q components are RFcomponents that are down-converted to baseband.

Step 1620 includes receiving a clock signal set according to a desiredoutput signal frequency of the desired output signal. In the example ofFIG. 15, step 1620 is achieved by receiving clock signal 1510.

Step 1630 includes processing the I and Q components to generatein-phase and quadrature amplitude information of first and secondconstant envelope constituent signals of the desired output signal. Inthe example of FIG. 15, the in-phase and quadrature amplitudeinformation is illustrated by U_(x), U_(y), L_(x), and L_(y).

Step 1640 includes processing the amplitude information and the clocksignal to generate the first and second constant envelope constituentsignals of the desired output signal. In an embodiment, the first andsecond constant envelope constituent signals are modulated according tothe desired output signal frequency. In the example of FIG. 15, step1640 is achieved by vector modulators 1520 and 1530, clock signal 1510,and amplitude information signals 1526, 1528, 1536, and 1538 to generatesignals 1540 and 1542.

Step 1650 includes amplifying the first and second constant envelopeconstituents, and summing the amplified signals to generate the desiredoutput signal. In an embodiment, the amplification of the first andsecond constant envelope constituents is according to a desired powerlevel of the desired output signal. In the example of FIG. 15, step 1650is achieved by PAs 1550 and 1560 amplifying respective signals 1540 and1542 and, subsequently, by the summing of amplified signals 1562 and1564 to generate output signal 1574.

FIG. 17 is a block diagram that illustrates an exemplary embodiment of avector power amplifier 1700 implementing the process flowchart 1600.Optional components are illustrated with dashed lines, although otherembodiments may have more or less optional components.

Referring to FIG. 17, in-phase (I) and quadrature (Q) information signal1710 is received by an I and Q Data Transfer Function module 1716. In anembodiment, I and Q Data Transfer Function module 1716 samples signal1710 according to a sample clock 1212. I and Q information signal 1710includes baseband I and Q information.

In an embodiment, I and Q Data Transfer Function module 1716 processesinformation signal 1710 to generate information signals 1720, 1722,1724, and 1726. The operation of I and Q Data Transfer Function module1716 is further described below in section 3.4.

Referring to FIG. 17, information signal 1720 includes vector modulator1750 quadrature amplitude information that is processed through DAC 1730to generate signal 1740. Information signal 1722 includes vectormodulator 1750 in-phase amplitude information that is processed throughDAC 1732 to generate signal 1742. Signals 1740 and 1742 are calculatedto generate a substantially constant envelope signal 1754. Withreference to FIG. 14, for example, information signals 1720 and 1722include the upper quadrature and in-phase components U_(y) and U_(x),respectively.

Still referring to FIG. 17, information signal 1726 includes vectormodulator 1752 quadrature amplitude information that is processedthrough DAC 1736 to generate signal 1746. Information signal 1724includes vector modulator 1752 in-phase amplitude information that isprocessed through DAC 1734 to generate signal 1744. Signals 1744 and1746 are calculated to generate a substantially constant envelope signal1756. With reference to FIG. 14, for example, information signals 1724and 1726 include the lower in-phase and quadrature components L_(x) andL_(y), respectively.

In the exemplary embodiment of FIG. 17, information signals 1720, 1722,1724 and 1726 are digital signals. Accordingly, each of signals 1720,1722, 1724 and 1726 is fed into a corresponding digital-to-analogconverter (DAC) 1730, 1732, 1734, and 1736. The resolution and samplerates of DACs 1730, 1732, 1734, and 1736 are selected according to thespecific desired signaling schemes. DACs 1730, 1732, 1734, and 1736 arecontrolled by DAC clock signals 1721, 1723, 1725, and 1727,respectively. DAC clock signals 1721, 1723, 1725, and 1727 may bederived from a same clock or may be independent of each other.

In other embodiments, information signals 1720, 1722, 1724 and 1726 aregenerated in analog format and no DACs are required.

Referring to FIG. 17, DACs 1730, 1732, 1734, and 1736 convert digitalinformation signals 1720, 1722, 1724, and 1726 into corresponding analogsignals, and input these analog signals into optional interpolationfilters 1731, 1733, 1735, and 1737, respectively. Interpolation filters1731, 1733, 1735, and 1737, which also serve as anti-aliasing filters,shape the DACs output signals to produce the desired output waveform.Interpolation filters 1731, 1733, 1735, and 1737 generate signals 1740,1742, 1744, and 1746, respectively.

Still referring to FIG. 17, signals 1740, 1742, 1744, and 1746 are inputinto vector modulators 1750 and 1752. Vector modulators 1750 and 1752generate first and second constant envelope constituents. In theembodiment of FIG. 17, channel clock 1714 is set according to a desiredoutput signal frequency to thereby establish the frequency of the outputsignal 1770.

Referring to FIG. 17, vector modulator 1750 combines signal 1740,multiplied with a 90° shifted version of channel clock signal 1714, andsignal 1742, multiplied with a 0° shifted version of channel clocksignal 1714, to generate output signal 1754. In parallel, vectormodulator 1752 combines signal 1746, multiplied with a 90° shiftedversion of channel clock signal 1714, and signal 1744, multiplied with a0° shifted version of channel clock signal 1714, to generate outputsignal 1756.

Output signals 1754 and 1756 represent constant envelope signals. A sumof output signals 1754 and 1756 results in a carrier signal having the Iand Q characteristics of the original baseband signal. In embodiments,to generate a desired power level at the output of vector poweramplifier 1700, signals 1754 and 1756 are amplified and then summed. Inthe embodiment of FIG. 17, for example, signals 1754 and 1756 are,respectively, input into corresponding power amplifiers (PAs) 1760 and1762. In an embodiment, PAs 1760 and 1762 include switching poweramplifiers. Autobias circuitry 1718 controls the bias of PAs 1760 and1762. In the embodiment of FIG. 17, for example, autobias circuitry 1718provides a bias voltage 1728 to PAs 1760 and 1762.

In an embodiment, PAs 1760 and 1762 apply equal or substantially equalpower amplification to respective constant envelope signals 1754 and1756. In an embodiment, the power amplification is set according to thedesired output power level. In other embodiments of vector poweramplifier 1700, PA drivers are additionally employed to provideadditional power amplification capability to the amplifier. In theembodiment of FIG. 17, for example, PA drivers 1774 and 1776 areoptionally added, respectively, between vector modulators 1750 and 1752and subsequent PAs 1760 and 1762.

Respective output signals 1764 and 1766 of PAs 1760 and 1762 aresubstantially constant envelope signals. In the embodiment of FIG. 17,output signals 1764 and 1766 are coupled together to generate outputsignal 1770 of vector power amplifier 1700. In embodiments, it is notedthat the outputs of PAs 1760 and 1762 are directly coupled. Directcoupling in this manner means that there is minimal or no resistive,inductive, or capacitive isolation between the outputs of PAs 1760 and1762. In other words, outputs of PAs 1760 and 1762 are coupled togetherwithout intervening components. Alternatively, in an embodiment, theoutputs of PAs 1760 and 1762 are coupled together indirectly throughinductances and/or capacitances that result in low or minimal impedanceconnections, and/or connections that result in minimal isolation andminimal power loss. Alternatively, outputs of PAs 1760 and 1762 arecoupled using well known combining techniques, such as Wilkinson, hybridcouplers, transformers, or known active combiners. In an embodiment, thePAs 1760 and 1762 provide integrated amplification and power combiningin a single operation. In an embodiment, one or more of the poweramplifiers and/or drivers described herein are implemented usingmultiple input, single output (MISO) power amplification techniques,examples of which are shown in FIGS. 17A, 17B, and 51A-H.

Output signal 1770 represents a signal having the desired I and Qcharacteristics of the baseband signal and the desired output powerlevel and frequency. In embodiments of vector power amplifier 1700, apull-up impedance 1778 is coupled between the output of vector poweramplifier 1700 and a power supply. In other embodiments, an impedancematching network 1780 is coupled at the output of vector power amplifier1700. Output stage embodiments according to power amplification methodsand systems of the present invention will be further described below insection 3.5.

In other embodiments of vector power amplifier 1700, process detectorsare employed to compensate for any process and/or temperature variationsin circuitry of the amplifier. In the exemplary embodiment of FIG. 17,for example, process detector 1712 is optionally added to monitorvariations in PA drivers 1774 and 1776.

FIG. 17A is a block diagram that illustrates another exemplaryembodiment 1700A of a vector power amplifier implementing processflowchart 1600. Optional components are illustrated with dashed lines,although other embodiments may have more or less optional components.Embodiment 1700A illustrates a multiple-input single-output (MISO)implementation of the amplifier of FIG. 17. In the embodiment of FIG.17A, constant envelope signals 1754 and 1756, output from vectormodulators 1750 and 1760, are input into MISO PA 1790. MISO PA 1790 is atwo-input single-output power amplifier. In an embodiment, MISO PA 1790include elements 1760, 1762, 1772, 1774, and 1776 as shown in theembodiment of FIG. 17, or functional equivalents thereof. In anotherembodiment, MISO PA 1790 may include other elements, such aspre-drivers, not shown in the embodiment of FIG. 17. Further, MISO PA1790 is not limited to being a two-input PA as shown in FIG. 17A. Inother embodiments, as will be described further below with reference toFIGS. 51A-H, PA 1790 can have any number of inputs.

In another embodiment of embodiment 1700, shown as embodiment 1700B ofFIG. 17B, optional Autobias circuitry 1218 generates separate biascontrol signals 1715, 1717, and 1719, corresponding to Bias A, Bias B,and Bias C, respectively. Signals 1715, 1717, and 1719 may or may not begenerated separately within Autobias circuitry 1718, but are outputseparately as shown. Further, signals 1715, 1717, and 1719 may or maynot be related as determined by the biasing required for the differentstages of MISO PA 1790.

FIG. 18 is a block diagram that illustrates another exemplary embodiment1800 of a vector power amplifier according to the Direct Cartesian2-Branch VPA embodiment of FIG. 16. Optional components are illustratedwith dashed lines, although other embodiments may have more or lessoptional components.

In the exemplary embodiment of FIG. 18, a DAC 1820 of sufficientresolution and sample rate replaces DACs 1730, 1732, 1734, and 1736 ofthe embodiment of FIG. 17. DAC 1820 is controlled by a DAC clock 1814.

DAC 1820 receives information signal 1810 from I and Q Data TransferFunction module 1716. Information signal 1810 includes identicalinformation content to signals 1720, 1722, 1724, and 1726 in theembodiment of FIG. 17.

DAC 1820 may output a single analog signal at a time. Accordingly, asample-and-hold architecture may be used as shown in FIG. 18.

In the embodiment of FIG. 18, DAC 1820 sequentially outputs analogsignals 1822, 1824, 1826, and 1828 to sample-and-hold circuits 1832,1834, 1836, and 1838, respectively. In an embodiment, DAC 1820 is ofsufficient resolution and sample rate to replace DACs 1720, 1722, 1724,and 1726 of the embodiment of FIG. 17. An output selector 1812determines which of output signals 1822, 1824, 1826, and 1828 areselected for output.

DAC1820's DAC clock signal 1814, output selector signal 1812, andsample-and-hold clocks 1830A-D, and 1840 are controlled by a controlmodule that can be independent or integrated into transfer functionmodule 1716.

In an embodiment, sample-and-hold circuits 1832, 1834, 1836, and 1838sample and hold their respective values and, according to a clock signal1830A-D, release the values to a second set of sample-and-hold circuits1842, 1844, 1846, and 1848. For example, S/H 1832 release's its value toS/H 1842 according to a received clock signal 1830A. In turn,sample-and-hold circuits 1842, 1844, 1846, and 1848 hold the receivedanalog values, and simultaneously release the values to interpolationfilters 1852, 1854, 1856, and 1858 according to a common clock signal1840.

In another embodiment, a single set of S/H circuitry that includes S/H1832, 1834, 1836, and 1838 can be employed. Accordingly, S/H circuits1832, 1834, 1836, and 1838 receive analog values from DAC 1820, and eachsamples and holds its received value according to independent clocks1830A-D. For example, S/H 1832 is controlled by clock 1830A, which maynot be synchronized with clock 1830B that controls S/H 1834. Forexample, DAC 1820 outputs signals 1822, 1824, 1826, and 1828 withappropriately selected analog values calculated by transfer functionmodule 1716 to S/H circuits 1832, 1834, 1836, and 1838 in order tocompensate for the time differences between clocks 1830A-D.

Other aspects of vector power amplifier 1800 correspond substantially tothose described above with respect to vector power amplifier 1700.

FIG. 18A is a block diagram that illustrates another exemplaryembodiment 1800A of a vector power amplifier according to the DirectCartesian 2-Branch VPA embodiment. Optional components are illustratedwith dashed lines, although in other embodiments more or less componentsmay be optional. Embodiment 1800A is a Multiple Input Single Output(MISO) implementation of embodiment 1800 of FIG. 18.

In the embodiment of FIG. 18A, constant envelope signals 1754 and 1756,output from vector modulators 1750 and 1752, are input into MISO PA1860. MISO PA 1860 is a two-input single-output power amplifier. In anembodiment, MISO PA 1860 includes elements 1744, 1746, 1760, 1762, and1772 as shown in the embodiment of FIG. 18, or functional equivalentsthereof. In another embodiment, MISO PA 1860 may include other elements,such as pre-drivers, not shown in the embodiment of FIG. 17. Further,MISO PA 1860 is not limited to being a two-input PA as shown in FIG.18A. In other embodiments as will be described further below withreference to FIGS. 51A-H, PA 1860 can have any number of inputs.

The embodiment of FIG. 18A further illustrates two different sample andhold architectures with a single or two levels of S/H circuitry asshown. The two implementations have been described above with respect toFIG. 18.

Other aspects of vector power amplifier 1800A are substantiallyequivalent to those described above with respect to vector poweramplifiers 1700 and 1800.

3.4) I and Q Data to Vector Modulator Transfer Functions

In some of the above described embodiments, I and Q data transferfunctions are provided to transform received I and Q data into amplitudeinformation inputs for subsequent stages of vector modulation andamplification. For example, in the embodiment of FIG. 17, I and Q DataTransfer Function module 1716 processes I and Q information signal 1710to generate in-phase and quadrature amplitude information signals 1720,1722, 1724, and 1726 of first and second constant envelope constituents1754 and 1756 of signal r(t). Subsequently, vector modulators 1750 and1752 utilize the generated amplitude information signals 1720, 1722,1724, and 1726 to create the first and second constant envelopeconstituent signals 1754 and 1756. Other examples include modules 710,712, and 1216 in FIGS. 7, 8, 12, and 13. These modules implementtransfer functions to transform I and/or Q data into amplitudeinformation inputs for subsequent stages of vector modulation andamplification.

According to the present invention, I and Q Data Transfer Functionmodules may be implemented using digital circuitry, analog circuitry,software, firmware or any combination thereof.

Several factors affect the actual implementation of a transfer functionaccording to the present invention, and vary from embodiment toembodiment. In one aspect, the selected VPA embodiment governs theamplitude information output of the transfer function and associatedmodule. It is apparent, for example, that I and Q Data Transfer Functionmodule 1216 of the CPCP 2-Branch VPA embodiment 1200 differs in outputthan I and Q Data Transfer Function module 1716 of the Direct Cartesian2-Branch VPA embodiment 1700.

In another aspect, the complexity of the transfer function variesaccording to the desired modulation scheme(s) that need to be supportedby the VPA implementation. For example, the sample clock, the DAC samplerate, and the DAC resolution are selected in accordance with theappropriate transfer function to construct the desired outputwaveform(s).

According to the present invention, transfer function embodiments may bedesigned to support one or more VPA embodiments with the ability toswitch between the supported embodiments as desired. Further, transferfunction embodiments and associated modules can be designed toaccommodate a plurality of modulation schemes. A person skilled in theart will appreciate, for example, that embodiments of the presentinvention may be designed to support a plurality of modulation schemes(individually or in combination) including, but not limited to, BPSK,QPSK, OQPSK, DPSK, CDMA, WCDMA, W-CDMA, GSM, EDGE, MPSK, MQAM, MSK,CPSK, PM, FM, OFDM, and multi-tone signals. In an embodiment, themodulation scheme(s) may be configurable and/or programmable via thetransfer function module.

3.4.1) Cartesian 4-Branch VPA Transfer Function

FIG. 19 is a process flowchart 1900 that illustrates an example I and Qtransfer function embodiment according to the Cartesian 4-Branch VPAembodiment. The process begins at step 1910, which includes receiving anin-phase data component and a quadrature data component. In theCartesian 4-Branch VPA embodiment of FIG. 7A, for example, this isillustrated by I Data Transfer Function module 710 receiving Iinformation signal 702, and Q Data Transfer Function module 712receiving Q information signal 704. It is noted that, in the embodimentof FIG. 7A, I and Q Data Transfer Function modules 710 and 712 areillustrated as separate components. In implementation, however, I and QData Transfer Function modules 710 and 712 may be separate or combinedinto a single module.

Step 1920 includes calculating a phase shift angle between first andsecond substantially equal and constant envelope constituents of the Icomponent. In parallel, step 1920 also includes calculating a phaseshift angle between first and second substantially equal and constantenvelope constituents of the Q component. As described above, the firstand second constant envelope constituents of the I components areappropriately phased relative to the I component. Similarly, the firstand second constant envelope constituents of the Q components areappropriately phased relative to the Q component. In the embodiment ofFIG. 7A, for example, step 1920 is performed by I and Q Data TransferFunction modules 710 and 712.

Step 1930 includes calculating in-phase and quadrature amplitudeinformation associated with the first and second constant envelopeconstituents of the I component. In parallel, step 1930 includescalculating in-phase and quadrature amplitude information associatedwith the first and second constant envelope constituents of the Qcomponent. In the embodiment of FIG. 7A, for example, step 1930 isperformed by and I and Q Data Transfer Function modules 710 and 712.

Step 1940 includes outputting the calculated amplitude information to asubsequent vector modulation stage. In the embodiment of FIG. 7A, forexample, I and Q Transfer Function modules 710 and 712 output amplitudeinformation signals 722, 724, 726, and 728 to vector modulators 760,762, 764, and 766 through DACs 730, 732, 734, and 736.

FIG. 20 is a block diagram that illustrates an exemplary embodiment 2000of a transfer function module, such as transfer function modules 710 and712 of FIG. 7A, implementing the process flowchart 1900. In the exampleof FIG. 20, transfer function module 2000 receives I and Q data signals2010 and 2012. In an embodiment, I and Q data signals 2010 and 2012represent I and Q data components of a baseband signal, such as signals702 and 704 in FIG. 7A.

Referring to FIG. 20, in an embodiment, transfer function module 2000samples I and Q data signals 2010 and 2012 according to a sampling clock2014. Sampled I and Q data signals are received by components 2020 and2022, respectively, of transfer function module 2000. Components 2020and 2022 measure, respectively, the magnitudes of the sampled I and Qdata signals. In an embodiment, components 2020 and 2022 are magnitudedetectors.

Components 2020 and 2022 output the measured I and Q magnitudeinformation to components 2030 and 2032, respectively, of transferfunction module 2000. In an embodiment, the measured I and Q magnitudeinformation is in the form of digital signals. Based on the I magnitudeinformation, component 2030 calculates a phase shift angle φ_(I) betweenfirst and second equal and constant or substantially equal and constantenvelope constituents of the sampled I signal. Similarly, based on the Qmagnitude information, component 2032 calculates phase shift angle φ_(Q)between a first and second equal and constant or substantially equal andconstant envelope constituents of the sampled Q signal. This operationshall now be further described.

In the embodiment of FIG. 20, φ_(I) and φ_(Q) are illustrated asfunctions ƒ(|{right arrow over (I)}|) and ƒ(|{right arrow over (Q)}|) ofthe I and Q magnitude signals. In embodiments, functions ƒ(|{right arrowover (I)}|) and ƒ(|{right arrow over (Q)}|) are set according to therelative magnitudes of the baseband I and Q signals respectively.ƒ(|{right arrow over (I)}|) and ƒ(|{right arrow over (Q)}|) according toembodiments of the present invention will be further described below insection 3.4.4.

Referring to FIG. 20, components 2030 and 2032 output the calculatedphase shift information to components 2040 and 2042, respectively. Basedon phase shift angle φ_(I), component 2040 calculates in-phase andquadrature amplitude information of the first and second constantenvelope constituents of the sampled I signal. Similarly, based on phaseshift angle φ_(Q), component 2042 calculates in-phase and quadratureamplitude information of the first and second constant envelopeconstituents of the sampled Q signal. Due to symmetry, in embodiments ofthe invention, calculation is required for 4 values only. In the exampleof FIG. 20, the values are illustrated as sgn(I)×I_(UX), I_(UY), Q_(UX),and sgn(Q)×Q_(UY), as provided in FIG. 5.

Components 2040 and 2042 output the calculated amplitude information tosubsequent stages of the vector power amplifier. In embodiments, each ofthe four calculated values is output separately to a digital-to-analogconverter. As shown in the embodiment of FIG. 7A for example, signals722, 724, 726, and 728 are output separately to DACs 730, 732, 734, and736, respectively. In other embodiments, signals 722, 724, 726, and 728are output into a single DAC as shown in FIGS. 800A and 800B.

3.4.2) CPCP 2-Branch VPA Transfer Function

FIG. 21 is a process flowchart 2100 that illustrates an example I and Qtransfer function embodiment according to the CPCP 2-Branch VPAembodiment. The process begins at step 2110, which includes receivingin-phase (I) and quadrature (Q) data components of a baseband signal. Inthe CPCP 2-Branch VPA embodiment of FIG. 12, for example, this isillustrated by I and Q Data Transfer Function module 1216 receiving Iand Q information signal 1210.

Step 2120 includes determining the magnitudes |I| and |Q| of thereceived I and Q data components.

Step 2130 includes calculating a magnitude |R| of the baseband signalbased on the measured |I| and |Q| magnitudes. In an embodiment, |R| issuch that |R|²=|I|²+|Q|². In the embodiment of FIG. 12, for example,steps 2120 and 2130 are performed by I and Q Data Transfer Functionmodule 1216 based on received information signal 1210.

Step 2140 includes normalizing the measured |I| and |Q| magnitudes. Inan embodiment, |I| and |Q| are normalized to generate an Iclk_phase andQclk_phase signals (as shown in FIG. 10) such that |I_(clk) _(—)_(phase)|²+|Q_(clk) _(—) _(phase)|²=constant. In the embodiment of FIG.12, for example, step 2140 is performed by I and Q Data TransferFunction module 1216 based on received information signal 1210.

Step 2150 includes calculating in-phase and quadrature amplitudeinformation associated with first and second constant envelopeconstituents. In the embodiment of FIG. 12, for example, step 2150 isperformed by I and Q Data Transfer Function module 1216 based on theenvelope magnitude |R|.

Step 2160 includes outputting the generated Iclk_phase and Qclk_phase(from step 2140) and the calculated amplitude information (from step2150) to appropriate vector modulators. In the embodiment of FIG. 12,for example, I and Q Data Transfer Function module 1216 outputinformation signals 1220, 1222, 1224, and 1226 to vector modulators1238, 1260, and 1262 through DACs 1230, 1232, 1234, and 1236.

FIG. 22 is a block diagram that illustrates an exemplary embodiment 2200of a transfer function module (such as module 1216 of FIG. 12)implementing the process flowchart 2100. In the example of FIG. 22,transfer function module 2200 receives I and Q data signal 2210. In anembodiment, I and Q data signal 2210 includes I and Q components of abaseband signal, such as signal 1210 in the embodiment of FIG. 12, forexample.

In an embodiment, transfer function module 2200 samples I and Q datasignal 2210 according to a sampling clock 2212. Sampled I and Q datasignals are received by component 2220 of transfer function module 2200.Component 2220 measures the magnitudes |{right arrow over (I)}| and|{right arrow over (Q)}| of the sampled I and Q data signals.

Based on the measured |{right arrow over (I)}| and |{right arrow over(Q)}| magnitudes, component 2230 calculates the magnitude |R| of thebaseband signal. In an embodiment, |{right arrow over (R)}| is such that|{right arrow over (R)}|²=|{right arrow over (I)}|²+|{right arrow over(Q)}|².

In parallel, component 2240 normalizes the measured |{right arrow over(I)}| and |{right arrow over (Q)}| magnitudes. In an embodiment, |{rightarrow over (I)}| and |{right arrow over (Q)}| are normalized to generateIclk_phase and Qclk_phase signals such that|Iclk_phase|²+|Qclk_phase|²=constant, where |Iclk_phase| and|Qclk_phase| represent normalized magnitudes of |{right arrow over (I)}|and |{right arrow over (Q)}|. Typically, given that the constant has avalue A, the measured |{right arrow over (I)}| and |{right arrow over(I)}| magnitudes are both divided by the quantity

$\frac{A}{\sqrt{{\overset{->}{I}}^{2} + {\overset{->}{Q}}^{2}}}$

Component 2250 receives the calculated |{right arrow over (R)}|magnitude from component 2230, and based on it calculates a phase shiftangle φ between first and second constant envelope constituents. Usingthe calculated phase shift angle φ, component 2050 then calculatesin-phase and quadrature amplitude information associated with the firstand second constant envelope constituents.

In the embodiment of FIG. 22, the phase shift angle φ is illustrated asa function f(|{right arrow over (R)}|) of the calculated magnitude|{right arrow over (R)}|.

Referring to FIG. 22, components 2240 and 2250 output the normalized|Iclk_phase| and |Qclk_phase| magnitude information and the calculatedamplitude information to DAC's for input into the appropriate vectormodulators. In embodiments, the output values are separately output todigital-to-analog converters. As shown in the embodiment of FIG. 12, forexample, signals 1220, 1222, 1224, and 1226 are output separately toDACs 1230, 1232, 1234, and 1236, respectively. In other embodiments,signals 1220, 1222, 1224, and 1226 are output into a single DAC as shownin FIGS. 13 and 13A.

3.4.3) Direct Cartesian 2-Branch Transfer Function

FIG. 23 is a process flowchart 2300 that illustrates an example I and Qtransfer function embodiment according to the Direct Cartesian 2-BranchVPA embodiment. The process begins at step 2310, which includesreceiving in-phase (I) and quadrature (Q) data components of a basebandsignal. In the Direct Cartesian 2-Branch VPA embodiment of FIG. 17, forexample, this is illustrated by I and Q Data Transfer Function module1716 receiving I and Q information signal 1710.

Step 2320 includes determining the magnitudes |{right arrow over (I)}|and |{right arrow over (Q)}| of the received I and Q data components.

Step 2330 includes calculating a magnitude |R| of the baseband signalbased on the measured |I| and |Q| magnitudes. In an embodiment, |R| issuch that |R|²=|I|²+|Q|². In the embodiment of FIG. 17, for example,steps 2320 and 2330 are performed by I and Q Data Transfer Functionmodule 1716 based on received information signal 1710.

Step 2340 includes calculating a phase shift angle θ of the basebandsignal based on the measured |I| and |Q| magnitudes. In an embodiment, θis such that

${\theta = {\tan^{- 1}\left( \frac{Q}{I} \right)}},$and wherein the sign of I and Q determine the quadrant of θ. In theembodiment of FIG. 17, for example, step 2340 is performed by I and QData Transfer Function module 1216 based on I and Q data componentsreceived in information signal 1210.

Step 2350 includes calculating in-phase and quadrature amplitudeinformation associated with a first and second constant envelopeconstituents of the baseband signal. In the embodiment of FIG. 17, forexample, step 2350 is performed by I and Q Data Transfer Function module1716 based on previously calculated magnitude |R| and phase shift angleθ.

Step 2360 includes outputting the calculated amplitude information toDAC's for input into the appropriate vector modulators. In theembodiment of FIG. 17, for example, I and Q Data Transfer Functionmodule 1716 output information signals 1720, 1722, 1724, and 1726 tovector modulators 1750 and 1752 through DACs 1730, 1732, 1734, and 1736.In other embodiments, signals 1720, 1722, 1724, and 1726 are output intoa single DAC as shown in FIGS. 18 and 18A.

FIG. 24 is a block diagram that illustrates an exemplary embodiment 2400of a transfer function module implementing the process flowchart 2300.In the example of FIG. 24, transfer function module 2400 (such astransfer function module 1716) receives I and Q data signal 2410, suchas signal 1710 in FIG. 17. In an embodiment, I and Q data signal 2410includes I and Q data components of a baseband signal.

In an embodiment, transfer function module 2400 samples I and Q datasignal 2410 according to a sampling clock 2412. Sampled I and Q datasignals are received by component 2420 of transfer function module 2200.Component 2420 measures the magnitudes |{right arrow over (I)}| and|{right arrow over (Q)}| of the sampled I and Q data signals.

Based on the measured |{right arrow over (I)}| and |{right arrow over(Q)}| magnitudes, component 2430 calculates the magnitude |{right arrowover (R)}|. In an embodiment, |{right arrow over (R)}| is such that|{right arrow over (R)}|²=|{right arrow over (I)}|²+|{right arrow over(Q)}|².

In parallel, component 2240 calculates the phase shift angle θ of thebaseband signal. In an embodiment, θ is such that

${\theta = {\tan^{- 1}\left( \frac{\overset{->}{Q}}{\overset{->}{I}} \right)}},$where the sign of I and Q determine the quadrant of θ.

Component 2450 receives the calculated |{right arrow over (R)}|magnitude from component 2430, and based on it calculates a phase shiftangle φ between first and second constant envelope constituent signals.In the embodiment of FIG. 24, the phase shift angle φ is illustrated asa function f₃(|{right arrow over (R)}|) of the calculated magnitude|{right arrow over (R)}|. This is further described in section 3.4.4.

In parallel, component 2450 receives the calculated phase shift angle θfrom component 2440. As functions of φ and θ, component 2450 thencalculates in-phase and quadrature amplitude information for the vectormodulator inputs that generate the first and second constant envelopeconstituents. In an embodiment, the in-phase and quadrature amplitudeinformation supplied to the vector modulators are according to theequations provided in (18).

Component 2450 outputs the calculated amplitude information tosubsequent stages of the vector power amplifier. In embodiments, theoutput values are separately output to digital-to-analog converters. Asshown in the embodiment of FIG. 17, for example, signals 1720, 1722,1724, and 1726 are output separately to DACs 1730, 1732, 1734, and 1736,respectively. In other embodiments, signals 1720, 1722, 1724, and 1726are output into a single DAC as shown in FIGS. 18 and 18A.

3.4.4) Magnitude to Phase Shift Transform

Embodiments of f(|I|), f(|Q|) of FIG. 20 and f(|R|) of FIGS. 22 and 24shall now be further described.

According to the present invention, any periodic waveform that can berepresented by a Fourier series and a Fourier transform can bedecomposed into two or more constant envelope signals.

Below are provided two examples for sinusoidal and square waveforms.

3.4.4.1) Magnitude to Phase Shift Transform for Sinusoidal Signals:

Consider a time-varying complex envelope sinusoidal signal r(t). In thetime domain, it can be represented as:r(t)=R(t) sin(ωt+δ(t))  (20)where R(t) represents the signal's envelope magnitude at time t, δ(t)represents the signal's phase shift angle at time t, and ω representsthe signal's frequency in radians per second.

It can be verified that, at any time instant t, signal r(t) can beobtained by the sum of two appropriately phased equal and constant orsubstantially equal and constant envelope signals. In other words, itcan be shown that:R(t)sin(ωt+δ(t))=A sin(ωt)+A sin(ωt+φ(t))  (21)for an appropriately chosen phase shift angle φ(t) between the twoconstant envelope signals. The phase shift angle φ(t) will be derived asa function of R(t) in the description below. This is equivalent to themagnitude to phase shift transform for sinusoidal signals.

Using a sine trigonometric identity, equation (21) can be re-written as:R(t)sin(ωt+δ(t))=A sin(ωt)+A sin(ωt)cos φ(t)+A sin(φ(t))cos ωt;

R(t)sin(ωt+δ(t))=A sin(φ(t))cos ωt+A(1+cos φ(t))sin ωt.  (22)

Note, from equation (22), that signal r(t) is written as a sum of anin-phase component and a quadrature component. Accordingly, the envelopemagnitude R(t) can be written as:R(t)=√{square root over ((A sin(φ(t)))²+(A(1+cos(φ(t))))²)}{square rootover ((A sin(φ(t)))²+(A(1+cos(φ(t))))²)};

R(t)=√{square root over (2A(A+cos(φ(t))).)}  (23)

Equation (23) relates the envelope magnitude R(t) of signal r(t) to thephase shift angle φ(t) between two constant envelope constituents ofsignal r(t). The constant envelope constituents have equal orsubstantially equal envelope magnitude A, which is typically normalizedto 1.

Inversely, from equation (23), the phase shift angle φ(t) can be writtenas a function of R(t) as follows:

$\begin{matrix}{{\phi(t)} = {{\arccos\left( {\frac{{R(t)}^{2}}{2A^{2}} - 1} \right)}.}} & (24)\end{matrix}$

Equation (24) represents the magnitude to phase shift transform for thecase of sinusoidal signals, and is illustrated in FIG. 26.

3.4.4.2) Magnitude to Phase Shift Transform for Square Wave Signals:

FIG. 28 illustrates a combination of two constant envelope square wavesignals according to embodiments of the present invention. In FIG. 28,signals 2810 and 2820 are constant envelope signals having a period T, aduty cycle γT (0<γ<1), and envelope magnitudes A1 and A2, respectively.

Signal 2830 results from combining signals 2810 and 2820. According toembodiments of the present invention, signal 2830 will have a magnitudeequal or substantially equal to a product of signals 2810 and 2820. Inother words, signal 2830 will have a magnitude of zero whenever eitherof signals 2810 or 2820 has a magnitude of zero, and a non-zeromagnitude when both signals 2810 and 2820 have non-zero magnitudes.

Further, signal 2830 represents a pulse-width-modulated signal. In otherwords, the envelope magnitude of signal 2830 is determined according tothe pulse width of signal 2830 over one period of the signal. Morespecifically, the envelope magnitude of signal 2830 is equal orsubstantially to the area under the curve of signal 2830.

Referring to FIG. 28, signals 2810 and 2820 are shown time-shiftedrelative to each other by a time shift t′. Equivalently, signals 2810and 2820 are phase-shifted relative to each other by a phase shift angle

$\phi = {\left( \frac{t^{\prime}}{T} \right) \times 2\pi}$radians.

Still referring to FIG. 28, note that the envelope magnitude R of signal2830, in FIG. 28, is given by:R=A ₁ ×A ₂×(γT−t′)  (25)

Accordingly, it can be deduced that φ is related to R according to:

$\begin{matrix}{\phi = {\left\lbrack {\gamma - \frac{R}{T\left( {A_{1}A_{2}} \right)}} \right\rbrack \times {\left( {2\pi} \right).}}} & (26)\end{matrix}$

Note, from equation (26), that R is at a maximum of γA1A2 when φ=0. Inother words, the envelope magnitude is at a maximum when the twoconstant envelope signals are in-phase with each other.

In typical implementations, signals 2810 and 2820 are normalized andhave equal or substantially equal envelope magnitude of 1. Further,signals 2810 and 2820 typically have a duty cycle of 0.5. Accordingly,equation (28) reduces to:

$\begin{matrix}{\phi = {\left\lbrack {0.5 - \frac{R}{T}} \right\rbrack \times {\left( {2\pi} \right).}}} & (27)\end{matrix}$

Equation (27) illustrates the magnitude to phase shift transform for thecase of normalized and equal or substantially equal envelope magnitudesquare wave signals. Equation (27) is illustrated in FIG. 26.

3.4.5) Waveform Distortion Compensation

In certain embodiments, magnitude to phase shift transforms may not beimplemented exactly as theoretically or practically derived. In fact,several factors may exist that require adjustment or tuning of thederived transform for optimal operation. Waveform distortioncompensation is one factor that may require adjustment or tuning of themagnitude to phase shift transform. Waveform distortion compensation isnow described below.

In practice, several factors may cause waveform distortion of constantenvelope constituents of a desired output signal r(t). Furthermore, itis expected that waveform distortion in the constituents translates intowaveform distortion in the desired output signal when the constituentsare combined. FIG. 25 illustrates the effect of waveform distortion on asignal using phasor signal representation. In FIG. 25, {right arrow over(R)} represents a phasor representation of a desired signal r(t). Inreality, waveform distortion can cause the actual phasor representationof r(t) to fall anywhere within a maximum error vector magnitude from{right arrow over (R)}. In the example of FIG. 25, this is illustratedusing a circle centered at {right arrow over (R)} and having a radiusequal or substantially equal to the maximum error vector magnitude.Phasors {right arrow over (R₁)} and {right arrow over (R₂)} representexamples of actual phasor representations of the desired signal r(t).

According to embodiments of the present invention, expected waveformdistortion to a desired output signal can be estimated. In someembodiments, the expected waveform distortion is compensated for at thevector modulation stage of the power amplifier. In other embodiments,the expected waveform distortion is compensated for at the transferfunction stage of the power amplifier.

In the former approach, compensation is achieved by applying appropriatemagnitude and phase shift tuning at the vector modulation stage outputs.Accordingly, waveform distortion is removed by waveshaping of theconstituent signals of the desired output signal.

In the latter approach, the transfer function is designed to factor inand cancel or at least reduce the effects of the expected waveformdistortion. As can be understood from the above description of transferfunctions, waveform distortion compensation can be introduced atdifferent positions within a transfer function stage. In someembodiments, compensation is applied at the output stage of the transferfunction. For example, referring to FIG. 17, transfer function moduleoutputs 1720, 1722, 1274 and 1726 may be tuned prior to being input intoDACs 1730, 1732, 1734, and 1736. In other embodiments, compensation isapplied at the magnitude to phase shift transform stage of the transferfunction. Referring to FIG. 24, for example, magnitude to phase shifttransform φ=ƒ(|R|) may be adjusted appropriately to compensate for theexpected waveform distortion.

3.5) Output Stage

An aspect of embodiments of the present invention lies in summingconstituent signals at the output stage of a vector power amplifier(VPA). This is shown, for example, in FIG. 7 where the outputs of PAs770, 772, 774, and 776 are summed. This is similarly shown in FIGS. 8,12, 13, 17, and 18, for example. Various embodiments for combining theoutputs of VPAs are described herein. While the following is describedin the context of VPAs, it should be understood that the followingteachings generally apply to coupling or summing the outputs of anyactive devices in any application.

FIG. 29 illustrates a vector power amplifier output stage embodiment2900 according to an embodiment of the present invention. Output stage2900 includes a plurality of vector modulator signals 2910-{1, . . . ,n} being input into a plurality of corresponding power amplifiers (PAs)2920-{1, . . . , n}. As described above, signals 2910-{1, . . . , n}represent constituent signals of a desired output signal of the vectorpower amplifier.

In the example of FIG. 29, PAs 2910-{1, . . . , n} equally amplify orsubstantially equally amplify input signals 2910-{1, . . . , n} togenerate amplified output signals 2930-{1, . . . , n}. Amplified outputsignals 2930-{1, . . . , n} are coupled together directly at summingnode 2940. According to this example embodiment of the presentinvention, summing node 2940 includes no coupling or isolating element,such as a power combiner, for example. In the embodiment of FIG. 29,summing node 2940 is a zero-impedance (or near-zero impedance)conducting wire. Accordingly, unlike in conventional systems that employcombining elements, the combining of output signals according to thisembodiment of the present invention incurs minimal power loss.

In another aspect, output stage embodiments of the present invention canbe implemented using multiple-input single-output (MISO) poweramplifiers.

In another aspect, output stage embodiments of the present invention canbe controlled to increase the power efficiency of the amplifier bycontrolling the output stage current according to the desired outputpower level.

In what follows, various output stage embodiments according to VPAembodiments of the present invention are provided in section 3.5.1. Insection 3.5.2, embodiments of output stage current shaping functions,for increasing the power efficiency of certain VPA embodiments of thepresent invention, are presented. Section 3.5.3 describes embodiments ofoutput stage protection techniques that may be utilized for certainoutput stage embodiments of the present invention.

3.5.1) Output Stage Embodiments

FIG. 30 is a block diagram that illustrates a power amplifier (PA)output stage embodiment 3000 according to an embodiment of the presentinvention. Output stage embodiment 3000 includes a plurality of PAbranches 3005-{1, . . . , n}. Signals 3010-{1, . . . , n} incoming fromrespective vector modulators represent inputs for output stage 3000.According to this embodiment of the present invention, signals 3010-{1,. . . , n} represent equal and constant or substantially equal andconstant envelope constituent signals of a desired output signal of thepower amplifier.

PA branches 3005-{1, . . . , n} apply equal or substantially equal poweramplification to respective signals 3010-{1, . . . , n}. In anembodiment, the power amplification level through PA branches 3005-{1, .. . , n} is set according to a power level requirement of the desiredoutput signal.

In the embodiment of FIG. 30, PA branches 3005-{1, . . . , n} eachincludes a power amplifier 3040-{1, . . . , n}. In other embodiments,drivers 3030-{1, . . . , n} and pre-drivers 3020-{1, . . . , n}, asillustrated in FIG. 30, may also be added in a PA branch prior to thepower amplifier element. In embodiments, drivers and pre-drivers areemployed whenever a required output power level may not be achieved in asingle amplifying stage.

To generate the desired output signal, outputs of PA branches 3005-{1, .. . , n} are coupled directly at summing node 3050. Summing node 3050provides little or no isolation between the coupled outputs. Further,summing node 3050 represents a relatively lossless summing node.Accordingly, minimal power loss is incurred in summing the outputs ofPAs 3040-{1, . . . , n}.

Output signal 3060 represents the desired output signal of output stage3000. In the embodiment of FIG. 30, output signal 3060 is measuredacross a load impedance 3070.

FIG. 31 is a block diagram that illustrates another power amplifier (PA)output stage embodiment 3100 according to the present invention. Similarto the embodiment of FIG. 30, output stage 3100 includes a plurality ofPA branches 3105-{1, . . . , n}. Each of PA branches 3105-{1, . . . , n}may include multiple power amplification stages represented by apre-driver 3020-{1, . . . , n}, driver 3030-{1, . . . , n}, and poweramplifier 3040-{1, . . . , n}. Output stage embodiment 3100 furtherincludes pull-up impedances coupled at the output of each poweramplification stage to provide biasing of that stage. For example,pull-up impedances 3125-{1, . . . , n} and 3135-{1, . . . , n},respectively, couple the pre-driver and driver stage outputs to powersupply or independent bias power supplies. Similarly, pull-up impedance3145 couples the PA stage outputs to the power supply or an independentbias power supply. According to this embodiment of the presentinvention, pull-up impedances represent optional components that mayaffect the efficiency but not necessarily the operation of the outputstage embodiment.

FIG. 32 is a block diagram that illustrates another power amplifier (PA)output stage embodiment 3200 according to the present invention. Similarto the embodiment of FIG. 30, output stage 3200 includes a plurality ofPA branches 3205-{1, . . . , n}. Each of PA branches 3205-{1, . . . , n}may include multiple power amplification stages represented by apre-driver 3020-{1, . . . , n}, driver 3030-{1, . . . , n}, and poweramplifier 3040-{1, . . . , n}. Output stage embodiment 3200 alsoincludes pull-up impedances coupled at the output of each poweramplification stage to achieve a proper biasing of that stage. Further,output stage embodiment 3200 includes matching impedances coupled at theoutputs of each power amplification stage to maximize power transferfrom that stage. For example, matching impedances 3210-{1, . . . , n}and 3220-{1, . . . , n}, are respectively coupled to the pre-driver anddriver stage outputs. Similarly, matching impedance 3240 is coupled atthe PA stage output. Note that matching impedance 3240 is coupled to thePA output stage subsequent to summing node 3250.

In the above-described embodiments of FIGS. 30-32, the PA stage outputsare combined by direct coupling at a summing node. For example, in theembodiment of FIG. 30, outputs of PA branches 3005-{1, . . . , n} arecoupled together at summing node 3050. Summing node 3050 is a nearzero-impedance conducting wire that provides minimal isolation betweenthe coupled outputs. Similar output stage coupling is shown in FIGS. 31and 32. It is noted that in certain embodiments of the presentinvention, output coupling, as shown in the embodiments of FIGS. 30-32or embodiments subsequently described below, may utilize certain outputstage protection measures. These protection measures may be implementedat different stages of the PA branch. Further, the type of protectionmeasures needed may be PA implementation-specific. A further discussionof output stage protection according to an embodiment of the presentinvention is provided in section 3.5.3.

FIG. 33 is a block diagram that illustrates another power amplifier (PA)output stage embodiment 3300 according to the present invention. Similarto the embodiment of FIG. 30, output stage 3300 includes a plurality ofPA branches 3305-{1, . . . , n}. Each of PA branches 3305-{1, . . . , n}may include multiple power amplification stages represented by apre-driver 3020-{1, . . . , n}, driver 3030-{1, . . . , n}, and poweramplifier 3040-{1, . . . , n}. Output stage embodiment 3300 may alsoinclude pull-up impedances 3125-{1, . . . , n}, 3135-{1, . . . , n}, and3145 coupled at the output of each power amplification stage to achievea proper biasing of that stage. Additionally, output stage embodiment3300 may include matching impedances 3210-{1, . . . , n}, 3220-{1, . . ., n}, and 3240 coupled at the output of each power amplification stageto maximize power transfer from that stage. Further, output stageembodiment 3300 receives an autobias signal 3310, from an Autobiasmodule 3340, coupled at the PA stage input of each PA branch 3305-{1, .. . , n}. Autobias module 3340 controls the bias of PAs 3040-{1, . . . ,n}. In an embodiment, autobias signal 3340 controls the amount ofcurrent flow through the PA stage according to a desired output powerlevel and signal envelope of the output waveform. A further descriptionof the operation of autobias signal and the autobias module is providedbelow in section 3.5.2.

FIG. 34 is a block diagram that illustrates another power amplifier (PA)output stage embodiment 3400 according to the present invention. Similarto the embodiment of FIG. 30, output stage 3400 includes a plurality ofPA branches 3405-{1, . . . , n}. Each of PA branches 3405-{1, . . . , n}may include multiple power amplification stages represented by apre-driver 3020-{1, . . . , n}, driver 3030-{1, . . . , n}, and poweramplifier 3040-{1, . . . , n}. Output stage embodiment 3400 may alsoinclude pull-impedances 3125-{1, . . . , n}, 3135-{1, . . . , n}, and3145 coupled at the output of each power amplification stage to achievedesired biasing of that stage. Additionally, output stage embodiment3400 may include matching impedances 3210-{1, . . . , n}, 3220-{1, . . ., n}, and 3240 coupled at the output of each power amplification stageto maximize power transfer from that stage. Further, output stageembodiment 3400 includes a plurality of harmonic control circuitnetworks 3410-{1, . . . , n} coupled at the PA stage input of each PAbranch {1, . . . , n}. Harmonic control circuit networks 3410-{1, . . ., n} may include a plurality of resistance, capacitance, and/orinductive elements and/or active devices coupled in series or inparallel. According to an embodiment of the present invention, harmoniccontrol circuit networks 3410-{1, . . . , n} provide harmonic controlfunctions for controlling the output frequency spectrum of the poweramplifier. In an embodiment, harmonic control circuit networks 3410-{1,. . . , n} are selected such that energy transfer to the fundamentalharmonic in the summed output spectrum is increased while the harmoniccontent of the output waveform is decreased. A further description ofharmonic control according to embodiments of the present invention isprovided below in section 3.6.

FIG. 35 is a block diagram that illustrates another power amplifier (PA)output stage embodiment 3500 according to the present invention. Outputstage embodiment 3500 represents a differential output equivalent ofoutput stage embodiment 3200 of FIG. 32. In embodiment 3500, PA stageoutputs 3510-{1, . . . , n} are combined successively to result in twoaggregate signals. The two aggregate signals are then combined across aloading impedance, thereby having the output of the power amplifierrepresent the difference between the two aggregate signals. Referring toFIG. 35, aggregate signals 3510 and 3520 are coupled across loadingimpedance 3530. The output of the power amplifier is measured across theloading impedance 3530 as the voltage difference between nodes 3540 and3550. According to embodiment 3500, the maximum output of the poweramplifier is obtained when the two aggregate signals are 180 degreesout-of-phase relative to each other. Inversely, the minimum output powerresults when the two aggregate signals are in-phase relative to eachother.

FIG. 36 is a block diagram that illustrates another output stageembodiment 3600 according to the present invention. Similar to theembodiment of FIG. 30, output stage 3600 includes a plurality of PAbranches 3605-{1, . . . , n}. Each of PA branches {1, . . . , n} mayinclude multiple power amplification stages represented by a pre-driver3020-{1, . . . , n}, a driver 3030-{1, . . . , n}, and a power amplifier(PA) 3620-{1, . . . , n}.

According to embodiment 3600, PA's 3620-{1, . . . , n} include switchingpower amplifiers. In the example of FIG. 36, power amplifiers 3620-{1, .. . , n} include npn bipolar junction transistor (BJT) elements Q1, . .. , Qn. BJT elements Q1, . . . , Qn have common collector nodes.Referring to FIG. 36, collector terminals of BJT elements Q1, . . . , Qnare coupled together to provide summing node 3640. Emitter terminals ofBJT elements Q1, . . . , Qn are coupled to a ground node, while baseterminals of BJT elements Q1, . . . , Qn provide input terminals intothe PA stage.

FIG. 37 is an example (related to FIG. 36) that illustrates an outputsignal of the PA stage of embodiment 3600 in response to square waveinput signals. For ease of illustration, a two-branch PA stage isconsidered. In the example of FIG. 37, square wave signals 3730 and 3740are input, respectively, into BJT elements 3710 and 3720. Note than wheneither of BJT elements 3710 or 3720 turns on, summing node 3750 isshorted to ground. Accordingly, when either of input signals 3730 or3740 is high, output signal 3780 will be zero. Further, output signal3780 will be high only when both input signals 3730 and 3740 are zero.According to this arrangement, PA stage 3700 performs pulse-widthmodulation, whereby the magnitude of the output signal is a function ofthe phase shift angle between the input signals.

Embodiments are not limited to npn BJT implementations as describedherein. A person skilled in the art will appreciate, for example, thatembodiments of the present invention may be implemented using pnp BJTs,CMOS, NMOS, PMOS, or other type of transistors. Further, embodiments canbe implemented using GaAs and/or SiGe transistors with the desiredtransistor switching speed being a factor to consider.

Referring back to FIG. 36, it is noted that while PAs 3620-{1, . . . ,n) are each illustrated using a single BJT notation, each PA 3620-{1, .. . , n} may include a plurality of series-coupled transistors. Inembodiments, the number of transistors included within each PA is setaccording to a required maximum output power level of the poweramplifier. In other embodiments, the number of transistors in the PA issuch that the numbers of transistors in the pre-driver, driver, and PAstages conform to a geometric progression.

FIG. 38 illustrates an exemplary PA embodiment 3800 according to anembodiment of the present invention. PA embodiment 3800 includes a BJTelement 3870, a LC network 3860, and a bias impedance 3850. BJT element3870 includes a plurality of BJT transistors Q1, . . . , Q8 coupled inseries. As illustrated in FIG. 38, BJT transistors Q1, . . . , Q8 arecoupled together at their base, collector, and emitter terminals.Collector terminal 3880 of BJT element 3870 provides an output terminalfor PA 3800. Emitter terminal 3890 of BJT element 3870 may be coupled tosubstrate or to an emitter terminal of a preceding amplifier stage. Forexample, emitter terminal 3890 is coupled to an emitter terminal of apreceding driver stage.

Referring to FIG. 38, LC network 3860 is coupled between PA inputterminal 3810 and input terminal 3820 of BJT element 3870. LC network3860 includes a plurality of capacitive and inductive elements.Optionally, a Harmonic Control Circuit network 3830 is also coupled atinput terminal 3820 of BJT element 3870. As described above, the HCCnetwork 3830 provides a harmonic control function for controlling theoutput frequency spectrum of the power amplifier.

Still referring to FIG. 38, bias impedance 3850 couples Iref signal 3840to input terminal 3820 of BJT element 3870. Iref signal 3840 representsan autobias signal that controls the bias of BJT element 3870 accordingto a desired output power level and signal envelope characteristics.

It is noted that, in the embodiment of FIG. 38, BJT element 3870 isillustrated to include 8 transistors. It can be appreciated by a personskilled in the art, however, that BJT element 3870 may include anynumber of transistors as required to achieve the desired output powerlevel of the power amplifier.

In another aspect, output stage embodiments can be implemented usingmultiple-input single-output (MISO) power amplifiers. FIG. 51A is ablock diagram that illustrates an exemplary MISO output stage embodiment5100A. Output stage embodiment 5100A includes a plurality of vectormodulator signals 5110-{1, . . . , n} that are input into MISO poweramplifier (PA) 5120. As described above, signals 5110-{1, . . . , n}represent constant envelope constituents of output signal 5130 of thepower amplifier. MISO PA 5120 is a multiple input single output poweramplifier. MISO PA 5120 receives and amplifies signals 5110-{1, . . . ,n} providing a distributed multi signal amplification process togenerate output signal 5130.

It is noted that MISO implementations, similar to the one shown in FIG.51A, can be similarly extended to any of the output stage embodimentsdescribed above. More specifically, any of the output stage embodimentsof FIGS. 29-37 can be implemented using a MISO approach. Additional MISOembodiments will now be provided with reference to FIGS. 51B-I. It isnoted that any of the embodiments described above can be implementedusing any of the MISO embodiments that will now be provided.

Referring to FIG. 51A, MISO PA 5120 can have any number of inputs asrequired by the substantially constant envelope decomposition of thecomplex envelope input signal. For example, in a two-dimensionaldecomposition, a two-input power amplifier can be used. According toembodiments of the present invention, building blocks for creating MISOPAs for any number of inputs are provided. FIG. 51B illustrates severalMISO building blocks according to an embodiment of the presentinvention. MISO PA 5110B represents a two-input single-output PA block.In an embodiment, MISO PA 5110B includes two PA branches. The PAbranches of MISO PA 5110B may be equivalent to any PA branches describedabove with reference to FIGS. 29-37, for example. MISO PA 5120Brepresents a three-input single-output PA block. In an embodiment, MISOPA 5120B includes three PA branches. The PA branches of MISO PA 5120Bmay equivalent to any PA branches described above with reference toFIGS. 29-37, for example.

Still referring to FIG. 51B, MISO PAs 5110B and 5120B represent basicbuilding blocks for any multiple-input single-output power amplifieraccording to embodiments of the present invention. For example, MISO PA5130B is a four-input single-output PA, which can be created by couplingtogether the outputs of two two-input single-output PA blocks, such asMISO PA 5110B, for example. This is illustrated in FIG. 51C. Similarly,it can be verified that MISO PA 5140B, an n-input single-output PA, canbe created from the basic building blocks 5110B and 5120B.

FIG. 51D illustrates various embodiments of the two-input single outputPA building block according to embodiments of the present invention.

Embodiment 5110D represents an npn implementation of the two-inputsingle output PA building block. Embodiment 5110D includes two npntransistors coupled together using a common collector node, whichprovides the output of the PA. A pull-up impedance (not shown) can becoupled between the common collector node and a supply node (not shown).

Embodiment 5130D represents a pnp equivalent of embodiment 5110D.Embodiment 5130D includes two pnp transistors coupled at a commoncollector node, which provides the output of the PA. A pull-downimpedance (not shown) can be coupled between the common collector nodeand a ground node (not shown).

Embodiment 5140D represents a complementary npn/pnp implementation ofthe two-input single output PA building block. Embodiment 5140D includesan npn transistor and a pnp transistor coupled at a common collectornode, which provides the output of the PA.

Still referring to FIG. 51D, embodiment 5120D represents a NMOSimplementation of the two-input single output PA building block.Embodiment 5120D includes two NMOS transistors coupled at a common drainnode, which provides the output of the PA.

Embodiment 5160D represents an PMOS equivalent of embodiment 5120D.Embodiment 5120D includes two PMOS transistors coupled at a common drainnode, which provides the output of the PA.

Embodiment 5150D represents a complementary MOS implementation of thetwo-input single-output PA building block. Embodiment 5150D includes aPMOS transistor and an NMOS transistor coupled at common drain node,which provides the output of the PA.

Two-input single-output embodiments of FIG. 51D can be further extendedto create multiple-input single-output PA embodiments. FIG. 51Eillustrates various embodiments of multiple-input single-output PAsaccording to embodiments of the present invention.

Embodiment 5150E represents an npn implementation of a multiple-inputsingle-output PA. Embodiment 5150E includes a plurality of npntransistors coupled together using a common collector node, whichprovides the output of the PA. A pull-up impedance (not shown) can becoupled between the common collector node and a supply voltage (notshown). Note that an n-input single-output PA according to embodiment5150E can be obtained by coupling additional npn transistors to thetwo-input single-output PA building block embodiment 5110D.

Embodiment 5170E represents a pnp equivalent of embodiment 5150E.Embodiment 5170E includes a plurality of pnp transistors coupledtogether using a common collector node, which provides the output of thePA. A pull-down impedance (not shown) may be coupled between the commoncollector node and a ground node (not shown). Note than an n-inputsingle-output PA according to embodiment 5170E can be obtained bycoupling additional pnp transistors to the two-input single-output PAbuilding block embodiment 5130D.

Embodiments 5110E and 5130E represent complementary npn/pnpimplementations of a multiple-input single-output PA. Embodiments 5110Eand 5130E may include a plurality of npn and/or pnp transistors coupledtogether using a common collector node, which provides the output of thePA. Note that an n-input single-output PA according to embodiment 5110Ecan be obtained by coupling additional npn and/or pnp transistors to thetwo-input single-output PA building block embodiment 5140D. Similarly,an n-input single-output PA according to embodiment 5130E can beobtained by coupling additional npn and/or pnp transistors to thetwo-input single-output PA building block embodiment 5130D.

Embodiment 5180E represents an PMOS implementation of a multiple-inputsingle-output PA. Embodiment 5180E includes a plurality of PMOStransistors coupled together using a common drain node, which providesthe output of the PA. Note that an n-input single-output PA according toembodiment 5180E can be obtained by coupling additional NMOS transistorsto the two-input single-output PA building block embodiment 5160D.

Embodiment 5160E represents a NMOS implementation of multiple-inputsingle-output PA. Embodiment 5160E includes a plurality of NMOStransistors coupled together using a common drain node, which providesthe output of the PA. Note that an n-input single-output PA according toembodiment 5160E can be obtained by coupling additional PMOS transistorsto the two-input single-output PA building block embodiment 5120D.

Embodiments 5120E and 5140E complementary MOS implementations of amultiple-input single-output PA. Embodiments 5120E and 5140E include aplurality of npn and pnp transistors coupled together using a commondrain node, which provides the output of the PA. Note that a n-inputsingle-output PA according to embodiment 5120E can be obtained bycoupling additional NMOS and/or PMOS transistors to the two-inputsingle-output PA building block 5150D. Similarly, an n-inputsingle-output PA according to embodiment 5140E can be obtained bycoupling additional NMOS and/or PMOS transistors to the two-inputsingle-output PA building block 5160D.

FIG. 51F illustrates further multiple-input single-output PA embodimentsaccording to embodiments of the present invention. Embodiment 5110Frepresents a complementary npn/pnp implementation of a multiple-inputsingle-output PA. Embodiment 5110F can be obtained by iterativelycoupling together embodiments of PA building block 5140D. Similarly,embodiment 5120F represents an equivalent NMOS/PMOS complementaryimplementation of a multiple-input single-output PA. Embodiment 5120Fcan be obtained by iteratively coupling together embodiments of PAbuilding block 5150D.

It must be noted that the multiple-input single-output embodimentsdescribed above may each correspond to a single or multiple branches ofa PA. For example, referring to FIG. 29, any of the multiple-inputsingle-output embodiments may be used to replace a single or multiplePAs 2920-{1, . . . , n}. In other words, each of PAs 2920-{1, . . . , n}may be implemented using any of the multiple-input single-output PAembodiments described above or with a single-input single-output PA asshown in FIG. 29.

It is further noted that the transistors shown in the embodiments ofFIGS. 51D, 51E, and 51F may each be implemented using a series oftransistors as shown in the exemplary embodiment of FIG. 38, forexample.

FIG. 51G illustrates further embodiments of the multiple-inputsingle-output PA building blocks. Embodiment 5110G illustrates anembodiment of the two-input single-output PA building block. Embodiment5110G includes two PA branches that can each be implemented according tosingle-input single-output or multiple-input single-output PAembodiments as described above. Further, embodiment 5110G illustrates anoptional bias control signal 5112G that is coupled to the two branchesof the PA embodiment. Bias control signal 5112G is optionally employedin embodiment 5110G based on the specific implementation of the PAbranches. In certain implementations, bias control will be required forproper operation of the PA. In other implementations, bias control isnot required for proper operation of the PA, but may provide improved PApower efficiency, output circuit protection, or power on currentprotection.

Still referring to FIG. 51G, embodiment 5120G illustrates an embodimentof the three-input single-output PA building block. Embodiment 5120Gincludes three PA branches that can each be implemented according tosingle-input single-output or multiple-input single-output PAembodiments as described above. Further, embodiment 5120G illustrates anoptional bias control signal 5114G that is coupled to the branches ofthe PA embodiment. Bias control signal 5114G is optionally employed inembodiment 5120G based on the specific implementation of the PAbranches. In certain implementations, bias control will be required forproper operation of the PA. In other implementations, bias control isnot required for proper operation of the PA, but may provide improved PApower efficiency.

FIG. 51H illustrates a further exemplary embodiment 5100H of thetwo-input single-output PA building block. Embodiment 5100H includes twoPA branches that can each be implemented according to single-inputsingle-output or multiple-input single-output PA embodiments asdescribed above. Embodiment 5100H further includes optional elements,illustrated using dashed lines in FIG. 51H, that can be additionallyemployed in embodiments of embodiment 5100H. In an embodiment, PAbuilding block 5100H may include a driver stage and/or pre-driver stagein each of the PA branches as shown in FIG. 51H. Process detectors mayalso be optionally employed to detect process and temperature variationsin the driver and/or pre-driver stages of the PA. Further, optional biascontrol may be provided to each of the pre-driver, driver, and/or PAstages of each branch of the PA embodiment. Bias control may be providedto one or more the stages based on the specific implementation of thatstage. Further, bias control may be required for certainimplementations, while it can be optionally employed in others.

FIG. 51I illustrates a further exemplary embodiment 5100I of amultiple-input single-output PA. Embodiment 5100I includes at least twoPA branches that can each be implemented according to single-inputsingle-output or multiple-input single-output PA embodiments asdescribed above. Embodiment 5100I further includes optional elementsthat can be additionally employed in embodiments of embodiment 5100I. Inan embodiment, the PA may include driver and/or pre-driver stages ineach of the PA branches as shown in FIG. 51I. Process detectors may alsobe optionally employed to detect process and temperature variations inthe driver and/or pre-driver stages of the PA. Further, optional biascontrol may be provided to each of the pre-driver, driver, and/or PAstages of each branch of the PA embodiment. Bias control may be providedto one or more the stages based on the specific implementation of thatstage. Further, bias control may be required for certainimplementations, while it can be optionally employed in others.

3.5.2) Output Stage Current Control—Autobias Module

Embodiments of the output stage and optional pre-driver and driver stagebias and current control techniques according to embodiments of thepresent invention are described below. In certain embodiments, outputstage current control functions are employed to increase the outputstage efficiency of a vector power amplifier (VPA) embodiment In otherembodiments, output stage current control is used to provide outputstage protection from excessive voltages and currents which is furtherdescribe in section 3.5.3. In embodiments, output stage current controlfunctions are performed using the Autobias module described above withreference to FIG. 33. A description of the operation of the Autobiasmodule in performing these current control functions is also presentedbelow according to an embodiment of the present invention.

According to embodiments of the present invention, power efficiency ofthe output stage of a VPA can be increased by controlling the outputstage current of the VPA as a function of the output power and theenvelope of the output waveform.

FIG. 37, illustrates a partial schematic of a Multiple Input SingleOutput amplifier comprised of two NPN transistors with input signals S1and S2. When S1 and S2 are designed to be substantially similarwaveforms and substantially constant envelope signals, any time varyingcomplex-envelope output signal can be created at circuit node 3750 bychanging the phase relationship of S1 and S2.

FIG. 39 illustrates an example time varying complex-envelope outputsignal 3910 and its corresponding envelope signal 3920. Note than signal3910 undergoes a reversal of phase at an instant of time t₀.Correspondingly, envelope signal 3920 undergoes a zero crossing at timet₀. Output signal 3910 exemplifies output signals according to typicalwireless signaling schemes such as W-CDMA, QPSK, and OFDM, for example.

FIG. 40 illustrates example diagram FIG. 37's output stage current inresponse to output signal 3910. I_(out) signal 4010 represents outputstage current without autobias control, and I_(out) signal 4020represents output stage current with autobias control. Without autobiascontrol, as the phase shift between S1 and S2 changes from 0 to 180degrees, the output current I_(out) increases. With autobias control,the output current I_(out) decreases and can be minimized when at ornear t₀ of FIG. 39.

Note that I_(out) signal 4020 varies as a function of envelope signal3920. Accordingly, I_(out) signal 4020 is at the maximum when a maximumoutput power is required, but decreases as the required output powergoes down. Particularly, I_(out) signal 4020 approaches zero as theassociated output power goes to zero. Accordingly, a person skilled inthe art will appreciate that output stage current control, according toembodiments of the present invention, results in significant powersavings and increases the power efficiency of the power amplifier.

According to embodiments of the present invention, output stage currentcontrol may be implemented according to a variety of functions. In anembodiment, the output stage current can be shaped to correspond to thedesired output power of the amplifier. In such an embodiment, the outputstage current is a function that is derived from the envelope of thedesired output signal, and the power efficiency will increase.

FIG. 41 illustrates exemplary autobias output stage current controlfunctions 4110 and 4120 according to embodiments of the presentinvention. Function 4110 may represent a function of output power andsignal envelope as described above. On the other hand, function 4120 mayrepresent a simple shaping function that goes to a minimum value for apre-determined amount of time when the output power is below a thresholdvalue. Accordingly, functions 4110 and 4120 represent two cases ofautobias output stage current control functions with autobias controlsignal 4110 resulting in I_(out) response 4130 and autobias controlsignal 4120 resulting in I_(out) response 4140. The invention, however,is not limited to those two exemplary embodiments. According toembodiments of the present invention, output stage autobias currentcontrol functions may be designed and implemented to accommodate theefficiency and current consumption requirements of a particular vectorpower amplifier design.

In implementation, several approaches exist for performing output stagecurrent control. In some embodiments, output stage current shaping isperformed using the Autobias module. The Autobias module is illustratedas autobias circuitry 714 and 716 in the embodiments of FIGS. 7 and 8.Similarly, the Autobias module is illustrated as autobias circuitry 1218in the embodiments of FIGS. 12 and 13, and as autobias circuitry 1718 inthe embodiments of FIGS. 17 and 18.

Output stage current control using Autobias is depicted in processflowchart 4800 of the embodiment of FIG. 48. The process begins in step4810, which includes receiving output power and output signal envelopeinformation of a desired output signal of a vector power amplifier(VPA). In some embodiments, implementing output stage current controlusing Autobias requires a priori knowledge of the desired output powerof the amplifier. Output power information may be in the form ofenvelope and phase information. For example, in the embodiments of FIGS.7, 8, 12, 13, 17, and 18, output power information is included in I andQ data components received by the VPA embodiment. In other embodiments,output power information may be received or calculated using othermeans.

Step 4820 includes calculating a signal according to the output powerand output envelope signal information. In embodiments, an Autobiassignal is calculated as a function of some measure of the desired outputpower. For example, the Autobias signal may be calculated as a functionof the envelope magnitude of the desired output signal. Referring to theembodiments of FIGS. 7, 8, 12, 13, 17, and 18, for example, it is notedthat the Autobias signal (signals 715 and 717 in FIGS. 7 and 8, signal1228 in FIGS. 12 and 13, and signals 1728 in FIGS. 17 and 18) iscalculated according to received I and Q data components of a desiredoutput signal. In certain embodiments, such as the ones described inFIGS. 7, 8, 12, 13, 17, and 18, the Autobias signal is calculated by anAutobias module being provided output power information. In otherembodiments, the Autobias signal may be calculated by the I and Q DataTransfer Function module(s) of the VPA. In such embodiments, an Autobiasmodule may not be required in implementation. In embodiments, the I andQ Data Transfer Function module calculates a signal, outputs the signalto a DAC which output signal represents the Autobias signal.

Step 4830 includes applying the calculated signal at an output stage ofthe VPA, thereby controlling a current of the output stage according tothe output power of the desired output signal. In embodiments, step 4830includes coupling the Autobias signal at the PA stage input of the VPA.This is illustrated, for example, in the embodiments of FIGS. 33 and 42where Autobias signal 3310 is coupled at the PA stage input of the VPAembodiment. In these embodiments, Autobias signal 3310 controls the biasof the PA stage transistors according to the output power of the desiredoutput signal of the VPA embodiment. For example, Autobias signal 3310may cause the PA stage transistors to operate in cutoff state when thedesired output power is minimal or near zero, thereby drawing little orno output stage current. Similarly, when a maximum output power isdesired, Autobias signal 3310 may bias the PA stage transistors tooperate in class C, D, E, etc. switching mode. Autobias signal 3310 mayalso cause the PA stage transistors or FETs to operate in forward orreverse biased states according to the desired output power and signalenvelope characteristics.

In other embodiments, step 4830 includes coupling the Autobias signalusing pull-up impedances at the PA stage input and optionally the inputsof the driver and pre-driver stages of the VPA. FIGS. 38 and 43illustrate such embodiments. For example, in the embodiment of FIG. 38,bias impedance 3850 couples Autobias Iref signal 3840 to input terminal3820 of BJT element 3870. BJT element 3870 represents the PA stage ofone PA branch of an exemplary VPA embodiment. Similarly, in theembodiment of FIG. 43, Autobias signal 4310 is coupled to transistorsQ1, . . . , Q8 through corresponding bias impedances Z1, . . . , Z8.Transistors Q1, . . . , Q8 represent the PA stage of one branch of anexemplary VPA embodiment.

Embodiments for implementing the Autobias circuitry described above willnow be provided. FIG. 27 illustrates three embodiments 2700A, 2700B, and2700C for implementing the Autobias circuitry. These embodiments areprovided for illustrative purposes, and are not limiting. Otherembodiments will be apparent to persons skilled in the art(s) based onthe teachings contained herein.

In embodiment 2700A, Autobias circuitry 2700A includes an AutobiasTransfer Function module 2712, a DAC 2714, and an optional interpolationfilter 2718. Autobias circuitry 2700A receives an I and Q Data signal2710. Autobias Transfer Function module 2712 processes the received Iand Q Data signal 2710 to generate an appropriate bias signal 2713.Autobias Transfer Function module 2712 outputs bias signal 2713 to DAC2714. DAC 2714 is controlled by a DAC clock 2716 which may be generatedin Autobias transfer module 2712. DAC 2714 converts bias signal 2713into an analog signal, and outputs the analog signal to interpolationfilter 2718. Interpolation filter 2718, which also serves as ananti-aliasing filter, shapes the DAC's output to generate Autobiassignal 2720, illustrated as Bias A in embodiment 5112G. Autobias signal2720 may be used to bias the PA stage and/or the driver stage, and/orthe pre-driver stage of the amplifier. In an embodiment, Autobias signal2720 may have several other Autobias signals derived therefrom to biasdifferent stages within the PA stage. This can be done using additionalcircuitry not included in embodiment 2700A.

In contrast, embodiment 2700B illustrates an Autobias circuitryembodiment in which multiple Autobias signals are derived within theAutobias circuitry. As shown in embodiment 2700B, circuit networks 2722,2726, and 2730, illustrated as circuit networks A, B, and C inembodiment 2700B, are used to derive Autobias signals 2724 and 2728 fromAutobias signal 2720. Autobias signals 2720, 2724, and 2728 are used tobias different amplification stages.

Embodiment 2700C illustrates another Autobias circuitry embodiment inwhich multiple Autobias signals are generated independently within theAutobias Transfer Function module 2712. In embodiment 2700C, AutobiasTransfer Function module 2712 generates multiple bias signals accordingto the received I and Q Data signal 2710. The bias signals may or maynot be related. Autobias Transfer Function module 2712 outputs thegenerated bias signals to subsequent DACs 2732, 2734, and 2736. DACs2732, 2734, and 2736 are controlled by DAC clock signals 2733, 2735, and2737, respectively. DACs 2732, 2734, and 2736 convert the received biassignals into analog signals, and output the analog signals to optionalinterpolation filters 2742, 2744, and 2746. Interpolation filters 2742,2744, and 2746, which also serve as anti-aliasing filters, shape theDACs outputs to generate Autobias signals 2720, 2724, and 2728. Similarto embodiment 2700B, Autobias signals 2720, 2724, and 2728 are used tobias different amplification stages such as the pre-driver, driver, andPA.

As noted above, Autobias circuitry embodiments according to the presentinvention are not limited to the ones described in embodiments 2700A,2700B, and 2700C. A person skilled in the art will appreciate, forexample, that Autobias circuitry can be extended to generate any numberof bias control signals as required to control the bias of variousstages of amplification, and not just three as shown in embodiments5200B and 5200C, for example.

3.5.3) Output Stage Protection

As described above, output stage embodiments according to embodiments ofthe present invention are highly power efficient as a result of beingable to directly couple outputs at the PA stage using no combining orisolating elements. Certain output stage embodiments in certaincircumstances and/or applications, however, may require additionalspecial output stage protection measures in order to withstand suchdirect coupling approach. This may be the case for example for outputstage embodiments such as 5110D, 5120D, 5130D, 5160D, 5150E, 5160E,5170E, and 5180E illustrated in FIGS. 51D and 51E. Note that, generally,complementary output stage embodiments, such as embodiments 5140D,5150D, 5110E, 5120E, 5130E, and 5140E of FIGS. 51D and 51E, do notrequire (but may optionally use) the same output stage protectionmeasures as will be described herein in this section. Output stageprotection measures and embodiments to support such measures are nowprovided.

In one aspect, transistors of distinct branches of a PA stage shouldgenerally not simultaneously be in opposite states of operation forextended periods of time. Following a restart or power on with no inputsbeing supplied to the final PA stages, transients within the PA branchesmay cause this mode to occur resulting in the PA stage transistorspotentially damaging one another or circuit elements connected to theoutput. Accordingly, embodiments of the present invention furtherconstrain the Autobias module to limit the output current in the PAstage.

In another aspect, it may be desired to ensure that the Autobias modulelimits the output voltages below the breakdown voltage specification ofthe PA stage transistors. Accordingly, in embodiments of the presentinvention, such as the one illustrated in FIG. 42 for example, afeedback element 4210 is coupled between the common collector node ofthe PA stage and the Autobias module. Feedback element 4210 monitors thecollector to base voltage of the PA stage transistors, and may constrainthe Autobias signal as necessary to protect the transistors and/orcircuit elements.

A person skilled in the art will appreciate that other output stageprotection techniques may also be implemented. Furthermore, output stageprotection techniques may be implementation specific. For example,depending on the type of PA stage transistors (npn, pnp, NMOS, PMOS,npn/pnp, NMOS/PMOS), different protection functions may be required.

3.6) Harmonic Control

According to embodiments of the present invention, an underlyingprinciple for each branch PA is to maximize the transfer of power to afundamental harmonic of the output spectrum. Typically, each branch PAmay be multi-stage giving rise to a harmonically rich output spectrum.In one aspect, transfer of real power is maximized for the fundamentalharmonic. In another aspect, for non-fundamental harmonics, real powertransfer is minimized while imaginary power transfer may be tolerated.Harmonic control, according to embodiments of the present invention, maybe performed in a variety of ways.

In one embodiment, real power transfer onto the fundamental harmonic ismaximized by means of wave-shaping of the PA stage input signals. Inpractice, several factors play a role in determining the optimal waveshape that results in a maximum real power transfer onto the fundamentalharmonic. Embodiment 3400 of the present invention, described above,represents one embodiment that employs waveshaping of PA stage inputsignals. In embodiment 3400, a plurality of harmonic control circuitry(HCC) networks 3410-{1, . . . , n} are coupled at the PA stage input ofeach PA branch {1, . . . , n}. HCC networks 3410-{1, . . . , n} have theeffect of waveshaping the PA stage inputs, and are typically selected soas to maximize real power transfer to the fundamental harmonic of thesummed output spectrum. According to embodiments of the presentinvention, waveshaping can be used to generate variations ofharmonically diverse waveforms. In other embodiments, as can be apparentto a person skilled in the art, waveshaping can be performed at thepre-driver and/or the driver stage.

In another embodiment, harmonic control is achieved by means ofwaveshaping of the PA stage output. FIG. 43 illustrates an exemplary PAstage embodiment 4300 of the present invention. In embodiment 4300,Autobias signal 4310 is coupled to transistors Q1, . . . , Q8 throughcorresponding bias impedances Z1, . . . , Z8. Notice that whenimpedances Z1, . . . , Z8 have different values, transistors Q1, . . . ,Q8 have different bias points and can be turned on at different times.This approach of biasing transistors Q1, . . . , Q8 is referred to asstaggered bias. Note that using staggered bias, the PA output waveformcan be shaped in a variety of ways depending on the values assigned tobias impedances Z1, . . . , Z8.

Harmonic control using staggered bias is depicted in process flowchart4900 of the embodiment of FIG. 49. The process begins in step 4910,which includes coupling an input signal at first ports of a plurality oftransistors of a power amplifier (PA) switching stage. In the exampleembodiment of FIG. 43, for example, step 4910 corresponds to couplingPA_IN signal 4310 at base terminals of the plurality of transistors Q1,. . . , Q8.

Step 4920 includes coupling a plurality of impedances between the firstports of the plurality of transistors and a bias signal. In the exampleembodiment of FIG. 43, for example, step 4920 is achieved by couplingimpedances Z1, . . . , Z8 between base terminals of respectivetransistors Q1, . . . , Q8 and Iref signal. In an embodiment, values ofthe plurality of impedances are selected to cause a time-staggeredswitching of the input signal, thereby harmonically shaping an outputsignal of the PA stage. In embodiments, a multi-stage staggered outputmay be generated by selecting multiple distinct values of the pluralityof impedances. In other embodiments, switching is achieved by selectingthe plurality of impedances to have equal or substantially equal value.

FIG. 44 illustrates an exemplary wave-shaped PA output using a two-stagestaggered bias approach. In a two-stage staggered bias approach, a firstset of the PA transistors is first turned on before a second set isturned on. In other words, the bias impedances take two differentvalues. Waveform 4410 represents an input waveform into the PA stage.Waveform 4420 represents the wave-shaped PA output according to atwo-stage staggered bias. Notice that output waveform 4420 slopes twiceas it transitions from 1 to 0, which corresponds to the first and secondsets of transistors turning on successively.

According to embodiments of the present invention, a variety ofmulti-stage staggered bias approaches may be designed. Bias impedancevalues may be fixed or variable. Furthermore, bias impedance values maybe equal or substantially equal, distinct, or set according to a varietyof permutations. For example, referring to the example of FIG. 43, oneexemplary permutation might set Z1=Z2=Z3=Z4 and Z5=Z6=Z7=Z8 resulting ina two-stage staggered bias.

3.7) Power Control

Vector power amplification embodiments of the present inventionintrinsically provide a mechanism for performing output power control.

FIG. 45 illustrates one approach for performing power control accordingto an embodiment of the present invention. In FIG. 45, phasors {rightarrow over (U₁)} and {right arrow over (L₁)} represent upper and lowerconstituents of a first phasor {right arrow over (R₁)}. {right arrowover (U₁)} and {right arrow over (L₁)} are constant magnitude and aresymmetrically shifted in phase relative to {right arrow over (R₁)} by aphase shift angle

$\frac{\phi}{2}.$Phasors {right arrow over (U₂)} and {right arrow over (L₂)} representupper and lower constituents of a second phasor {right arrow over (R₂)}.{right arrow over (U₂)} and {right arrow over (L₂)} are constantmagnitude and are symmetrically shifted in phase relative to {rightarrow over (R₂)} by a phase shift angle

$\frac{\phi}{2} + {\phi_{off}.}$

It is noted, from FIG. 45, that {right arrow over (R₁)} and {right arrowover (R₂)} are in-phase relative to each other but only differ inmagnitude. Furthermore, {right arrow over (U₂)} and {right arrow over(L₂)} are equally or substantially equally phased shifted relative to{right arrow over (U₁)} and {right arrow over (L₁)}, respectively.Accordingly, it can be inferred that, according to the presentinvention, a signal's magnitude can be manipulated without varying itsphase shift angle by equally or substantially equally shiftingsymmetrically its constituent signals.

According to the above observation, output power control can beperformed by imposing constraints on the phase shift angle of theconstituent signals of a desired output signal. Referring to FIG. 45,for example, by constraining the range of values that phase shift angle

$\frac{\phi}{2}$can take, magnitude constraints can be imposed on phasor {right arrowover (R₁)}.

According to embodiments of the present invention, a maximum outputpower level can be achieved by imposing a minimum phase shift anglecondition. For example, referring to FIG. 45, by setting a conditionsuch that

${\frac{\phi}{2} \geq \phi_{ff}},$the magnitude of phasor {right arrow over (R₁)} is constrained not toexceed a certain maximum level. Similarly, a maximum phase shift anglecondition imposes a minimum magnitude level requirement.

In another aspect of power control, output power resolution is definedin terms of a minimum power increment or decrement step size. Accordingto an embodiment of the present invention, output power resolution maybe implemented by defining a minimum phase shift angle step size.Accordingly, phase shift angle values are set according to a discretevalue range having a pre-determined step size. FIG. 46 illustrates anexemplary phase shift angle spectrum, whereby phase shift angle

$\frac{\phi}{2}$is set according to a pre-determined value range having a minimum stepφ_(step).

A person skilled in the art will appreciate that a variety of powercontrol schemes may be implemented in a fashion similar to thetechniques described above. In other words, various power controlalgorithms can be designed, according to the present invention, bysetting corresponding constraints on phase shift angle values. It isalso apparent, based on the description above of data transferfunctions, that power control schemes can be naturally incorporated intoa transfer function implementation.

3.8) Exemplary Vector Power Amplifier Embodiment

FIG. 47 illustrates an exemplary embodiment 4700 of a vector poweramplifier according to the present invention. Embodiment 4700 isimplemented according to the Direct Cartesian 2-Branch VPA method.

Referring to FIG. 47, signals 4710 and 4712 represent incoming signalsfrom a transfer function stage. The transfer function stage is not shownin FIG. 47. Block 4720 represents a quadrature generator which may beoptionally implemented according to an embodiment of the presentinvention. Quadrature generator 4720 generates clock signals 4730 and4732 to be used by vector modulators 4740 and 4742, respectively.Similarly, signals 4710 and 4712 are input into vector modulators 4740and 4742. As described above, vector modulators 4740 and 4742 generateconstant envelope constituents that are, subsequently, processed by a PAstage. In embodiment 4700, the PA stage is multi-stage, whereby each PAbranch includes a pre-driver stage 4750-4752, a driver stage 4760-4762,and a power amplifier stage 4770-4772.

Further illustrated in FIG. 47 are Autobias signals 4774 and 4776, andterminals 4780 and 4782 for coupling harmonic control circuitry andnetworks. Terminal node 4780 represents the output terminal of thevector power amplifier, and is obtained by direct coupling of the two PAbranches' outputs.

4. Summary

Mathematical basis for a new concept related to processing signals toprovide power amplification and up-conversion is provided herein. Thesenew concepts permit arbitrary waveforms to be constructed from sums ofwaveforms which are substantially constant envelope in nature. Desiredoutput signals and waveforms may be constructed from substantiallyconstant envelope constituent signals which can be created from theknowledge of the complex envelope of the desired output signal.Constituent signals are summed using new, unique, and novel techniquesnot available commercially, not taught or found in literature or relatedart. Furthermore, the blend of various techniques and circuits providedin the disclosure provide unique aspects of the invention which permitssuperior linearity, power added efficiency, monolithic implementationand low cost when compared to current offerings. In addition,embodiments of the invention are inherently less sensitive to processand temperature variations. Certain embodiments include the use ofmultiple input single output amplifiers described herein.

Embodiments of the invention can be implemented by a blend of hardware,software and firmware. Both digital and analog techniques can be usedwith or without microprocessors and DSP's.

Embodiments of the invention can be implemented for communicationssystems and electronics in general. In addition, and without limitation,mechanics, electro mechanics, electro optics, and fluid mechanics canmake use of the same principles for efficiently amplifying andtransducing signals.

5. Conclusion

The present invention has been described above with the aid offunctional building blocks illustrating the performance of specifiedfunctions and relationships thereof. The boundaries of these functionalbuilding blocks have been arbitrarily defined herein for the convenienceof the description. Alternate boundaries can be defined so long as thespecified functions and relationships thereof are appropriatelyperformed. Any such alternate boundaries are thus within the scope andspirit of the claimed invention. One skilled in the art will recognizethat these functional building blocks can be implemented by discretecomponents, application specific integrated circuits, processorsexecuting appropriate software and the like and combinations thereof.

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample only, and not limitation. Thus, the breadth and scope of thepresent invention should not be limited by any of the above-describedexemplary embodiments, but should be defined only in accordance with thefollowing claims and their equivalents.

What is claimed is:
 1. A method of power amplification, comprising:receiving information; receiving a frequency reference signal inaccordance with a desired output signal frequency; determining a complexenvelope from said received information; determining a phase shift angleusing said complex envelope, wherein said phase shift angle is equal toa phase difference between first and second substantially constantenvelope signals; using said phase shift angle to generate in-phase andquadrature phase amplitude control signals; controlling a phase angle ofsaid frequency reference signal to generate a phase angle controlledfrequency reference signal of equal phase as a desired output signal;generating said first and second substantially constant envelope signalsusing said in-phase and quadrature phase amplitude control signals andsaid phase angle controlled frequency reference signal; providing saidfirst and second substantially constant envelope signals to a MISO(multiple input single output) amplifier comprising two or more circuitbranches each comprising one or more transistors, wherein said first andsecond substantially constant envelope signals are amplified andcombined to create said desired output signal using said MISO amplifier.2. The method according to claim 1, wherein said step of receivinginformation comprises: receiving one or more baseband informationsignals.
 3. The method according to claim 1, wherein said step ofreceiving information comprises: receiving in-phase and quadrature phasebaseband information signals.
 4. The method according to claim 1,wherein said step of receiving information comprises: receiving analogin-phase and analog quadrature phase baseband information signals. 5.The method according to claim 1, wherein said step of receivinginformation comprises: receiving digital in-phase and digital quadraturephase baseband information signals.
 6. The method according to claim 1,wherein said step of receiving information comprises: receiving atime-varying radio frequency (RF) signal.
 7. The method according toclaim 1, wherein said step of receiving information comprises: receivingin-phase and quadrature phase radio frequency (RF) signals.
 8. Themethod according to claim 1, wherein said controlling step comprises:generating a plurality of phase control signals from said receivedinformation; generating a plurality of secondary frequency referencesignals from said frequency reference signal, wherein one or more ofsaid secondary frequency reference signals are phase shifted relative tosaid frequency reference signal; multiplying said secondary frequencyreference signals by respective said phase control signals, therebygenerating controlled secondary frequency reference signals; andcombining said controlled secondary frequency reference signals togenerate said phase angle controlled frequency reference signal.
 9. Themethod of claim 8, wherein said step of generating a plurality of phasecontrol signals comprises generating in-phase and quadrature phase phasecontrol signals from in-phase and quadrature phase componentsrespectively of the received information.
 10. The method of claim 9,wherein said step of generating secondary frequency reference signalscomprises generating in-phase and quadrature phase secondary frequencyreference signals relative to said frequency signal.
 11. The method ofclaim 10, wherein said multiplying step comprises multiplying thein-phase phase control signal by the in-phase secondary frequencyreference signal and the quadrature phase phase control signal by thequadrature phase secondary reference signal.
 12. The method of claim 9,wherein a phase angle of said phase angle controlled frequency referencesignal relative to a reference signal is according to a ratio of thein-phase and quadrature phase components of the received information.13. The method of claim 9, wherein a magnitude of said phase anglecontrolled frequency reference signal is according to:R_clk²=Iclk_phase²+Qclk_phase² wherein R_clk denotes the magnitude ofthe phase angle controlled frequency reference signal, I clk_phasedenotes a magnitude of the in-phase phase control signal, and Qclk_phase denotes a magnitude of the quadrature phase phase controlsignal.
 14. The method according to claim 1, wherein said generatingstep further comprises: generating ternary frequency reference signalsfrom said phase angle controlled frequency reference signal, wherein oneor more of said ternary frequency reference signals are phase shiftedrelative to said phase angle controlled frequency reference signal;multiplying said ternary frequency reference signals by respective saidamplitude control signals, thereby controlling said ternary frequencyreference signals; and combining subsets of said controlled ternaryfrequency reference signals to generate said first and secondsubstantially constant envelope signals.
 15. The method according toclaim 14, further comprising: converting from digital to analog saidgenerated in-phase and quadrature phase amplitude control signals. 16.The method according to claim 14, further comprising: interpolationfiltering said generated in-phase and quadrature phase amplitude controlsignals.
 17. The method according to claim 14, wherein said step ofgenerating a plurality of ternary frequency reference signals comprisesgenerating in-phase and quadrature phase ternary frequency referencesignals relative to said phase angle controlled frequency referencesignal.
 18. The method according to claim 17, wherein said multiplyingstep comprises multiplying the in-phase ternary frequency referencesignals by the in-phase amplitude control signals and the quadraturephase ternary frequency reference signals by the quadrature phaseamplitude control signals.
 19. The method of claim 17, said combiningsubsets step comprises combining the controlled ternary frequencyreference signals of the first substantially constant envelope signal togenerate the first substantially constant envelope signal, and combiningthe controlled ternary frequency reference signals of the secondsubstantially constant envelope signal to generate the secondsubstantially constant envelope signal.
 20. The method according toclaim 14, further comprising: generating said in-phase and quadraturephase amplitude control signals from said received information andtransfer functions, said transfer functions having been generated so asto enable said amplification and combination of said first and secondsubstantially constant envelope signals.
 21. The method of claim 20,wherein said transfer functions having been generated to compensate forany resistive, inductive, and capacitive elements between outputs ofsaid transistors in said circuit branches of said MISO amplifier. 22.The method of claim 20, wherein said transfer functions having beengenerated to compensate for any parasitic resistive, inductive, andcapacitive elements between outputs of said transistors in said circuitbranches of said MISO amplifier.
 23. The method of claim 20, saidtransfer functions having been generated to compensate for anyresistive, inductive, and capacitive components between outputs of saidtransistors in said circuit branches of said MISO amplifier.
 24. Themethod according to claim 1, wherein said phase shift angle isdetermined according to:$\phi = {{arc}\;{\cos\left( {\frac{R^{2}}{2A^{2}} - 1} \right)}}$wherein (φ) denotes said phase shift angle, R denotes said complexenvelope, and A denotes an optional constant.
 25. The method accordingto claim 1, wherein said in-phase and quadrature phase amplitude controlsignals are generated according to:${C = {A\;{\cos\left( \frac{\phi}{2} \right)}}};$${\alpha = {{- \beta} = {A\;{\sin\left( \frac{\phi}{2} \right)}}}};$wherein C denotes said in-phase amplitude control signals of the firstand second substantially constant envelope signals, α and β denoterespectively said quadrature phase amplitude control signals of thefirst and second substantially constant envelope signals, φ denotes saidphase shift angle, and A denotes an optional constant.
 26. The methodaccording to claim 1, wherein said phase shift angle is determinedaccording to a function mathematically equivalent to:$\phi = {{arc}\;{\cos\left( {\frac{R^{2}}{2A^{2}} - 1} \right)}}$wherein φ denotes said phase shift angle, R denotes said complexenvelope, and A denotes an optional constant.
 27. The method accordingto claim 1, wherein said in-phase and quadrature phase amplitude controlsignals are generated according to functions mathematically equivalentto: ${C = {A\;{\cos\left( \frac{\phi}{2} \right)}}};$${\alpha = {{- \beta} = {A\;{\sin\left( \frac{\phi}{2} \right)}}}};$wherein C denotes said in-phase amplitude control signals of the firstand second substantially constant envelope signals, α and β denoterespectively said quadrature phase amplitude control signals of thefirst and second substantially constant envelope signals, φ denotes saidphase shift angle, and A denotes an optional constant.
 28. The methodaccording to claim 1, further comprising: controlling biasing of one ormore power amplification stages of said MISO amplifier according to saidreceived information.
 29. The method according to claim 28, wherein saidcontrolling biasing step further comprises: using a bias controlfunction to control one or more of output power efficiency, outputlinearity, and amplification stage protection.
 30. The method accordingto claim 1, further comprising: non-linearly power amplifying said firstand second substantially constant envelope signals using said MISOamplifier.
 31. The method according to claim 1, further comprising:harmonically shaping said output signal.
 32. The method according toclaim 1, wherein said first and second substantially constant envelopesignals are combined using one or more biasing control functions. 33.The method according to claim 1, wherein said step of providing saidfirst and second substantially constant envelope signals to said MISOamplifier comprises: dynamically controlling said transistors of saidcircuit branches to thereby transition operation of said transistorsalong an operational range that includes switching amplifier operationand current source amplifier operation.
 34. The method according toclaim 1, wherein said step of providing said first and secondsubstantially constant envelope signals to said MISO amplifiercomprises: simultaneously controlling an input differential phase and abias of said transistors based on requirements of said desired outputsignal.
 35. The method according to claim 1, further comprising using abias control function to control one or more of output power efficiency,output linearity, and amplification stage protection of said MISO poweramplifier.
 36. The method of claim 1, further comprising: dynamicallycontrolling said transistors of said circuit branches to therebytransition operation of said transistors along an operational range thatincludes switching amplifier operation and current source amplifieroperation.
 37. The method of claim 1, wherein said substantiallyconstant envelope signals are amplified and combined with only parasiticresistive, inductive, or capacitive elements between outputs of saidtransistors in said circuit branches of said MISO amplifier.
 38. Themethod of claim 1, wherein said substantially constant envelope signalsare amplified and combined with substantially no electrical isolationbetween outputs of said transistors in said circuit branches of saidMISO amplifier.
 39. The method of claim 1, wherein output voltages ofsaid transistors in said circuit branches of said MISO amplifier aresubstantially equal to an output voltage of said MISO amplifier for alloperational instances of time of said MISO amplifier.